Method and apparatus for demodulating signals in a pulse oximetry system

ABSTRACT

A method and an apparatus measure blood oxygenation in a subject. A first signal source applies a first input signal during a first time interval. A second signal source applies a second input signal during a second time interval. A detector detects a first parametric signal responsive to the first input signal passing through a portion of the subject having blood therein. The detector also detects a second parametric signal responsive to the second input signal passing through the portion of the subject. The detector generates a detector output signal responsive to the first and second parametric signals. A signal processor receives the detector output signal and demodulates the detector output signal by applying a first demodulation signal to a signal responsive to the detector output signal to generate a first output signal responsive to the first parametric signal. The signal processor applies a second demodulation signal to the signal responsive to the detector output signal to generate a second output signal responsive to the second parametric signal. The first demodulation signal and the second demodulation signal both include at least a first component having a first frequency and a first amplitude and a second component having a second frequency and a second amplitude. The second frequency is a harmonic of the first frequency. The second amplitude is related to the first amplitude to minimize crosstalk from the first parametric signal to the second output signal and to minimize crosstalk from the second parametric signal to the first output signal.

REFERENCE TO PRIOR RELATED APPLICATION

[0001] This application is a divisional of U.S. application Ser. No.09/735,960 (which will issue as U.S. Pat. No. 6,643,530) filed Dec. 13,2000, which is a divisional of U.S. application Ser. No. 09/058,799 (nowU.S. Pat. No. 6,229,056) filed Apr. 10, 1998, which is acontinuation-in-part of U.S. application Ser. No. 09/005,898 (now U.S.Pat. No. 5,919,134) filed Jan. 12, 1998 which claims priority from U.S.Provisional Application No. 60/043,620 filed Apr. 14, 1997.

BACKGROUND OF THE INVENTION

[0002] 1. Field of the Invention

[0003] The present invention relates to the field of signal processing,and, more particularly, relates to the field of processing of signalsgenerated in a physiological monitoring system, such as, for example, ina system for measuring blood oxygen saturation using pulse oximetry.

[0004] 2. Description of the Related Art

[0005] The present invention will be described herein in connection witha pulse oximetry apparatus and a method, which are used to measure bloodoxygen saturation in a subject, such as, for example, a human patient.The teachings of the present invention can be used in other applicationswherein useable signal information is obtained in a noisy environment.

[0006] In an exemplary pulse oximetry apparatus and a correspondingmethod, blood oxygen saturation is determined by transmitting pulses ofelectromagnetic energy through a portion of a subject which has bloodflowing therein (e.g., through a finger, through an ear lobe, or otherportion of the body where blood flows close to the skin). In theexamples described herein, the pulses of electromagnetic energy compriseperiodic pulses of red light having wavelengths of approximately 660nanometers, for example, and periodic pulses of infrared light havingwavelengths of approximately 905 nanometers. As described, for example,in U.S. Pat. No. 5,482,036 and in U.S. Pat. No. 5,490,505, the pulses ofred light and the pulses of infrared light are applied with the sameperiodicity but in an alternating and non-overlapping manner. Inparticular, in preferred embodiments, the red pulses are active forapproximately 25% of each cycle and the infrared pulses are also activefor approximately 25% of each cycle. The red pulses are separated intime from the infrared pulses such that both pulses are inactive forapproximately 25% of each cycle between a red pulse and the nextinfrared pulse and both pulses are inactive for approximately 25% ofeach cycle between an infrared pulse and the next red pulse. (Althoughdescribed herein below in connection with pulses having 25% duty cycles,it should be understood by persons of skill in the art that the dutycycles of the pulses can be changed in some applications.) Afterpropagating through the portion of the subject, the red pulses and theinfrared pulses are detected by a detector which is responsive to lightat both wavelengths and which generates an electrical signal which has apredictable relationship to the intensity of the electromagnetic energyincident on the detector. The electrical signal is processed inaccordance with the present invention to provide a representation of theblood oxygen saturation of the subject.

[0007] In conventional time division multiplexing (TDM) demodulationthat uses rectangular waves to drive the red and infrared LEDs, theconventional process of demodulation using square waves can result inthe aliasing of the ambient noise components that come close to thesidebands of harmonics and the fundamental frequency of the rectangularwaves, and the noise components are thus collapsed into the outputsignal generated by the demodulation. In particular, it is verydifficult to avoid including harmonics of the line frequency in thedemodulated output signal.

SUMMARY OF THE INVENTION

[0008] The present invention avoids the problems associated withconventional demodulation and separation of TDM signals. In particular,the present invention avoids the problem of aliasing of the ambientnoise into the passband of the system by selectively demodulatingcertain harmonics of the TDM signal. For example, in one embodiment,only two harmonics (e.g., the fundamental and the first harmonic) aredemodulated. In other embodiments, more harmonics are demodulated. Thepresent invention specifically addresses solutions to problems caused bycrosstalk resulting from filtering and also resulting from demodulatingwith only certain harmonics instead of demodulating with all harmonicsas is done using conventional square wave demodulation. In a digitalimplementation of the present invention, the output of the photodetectoris initially sampled at a very high frequency (e.g., 46,875 Hz), and thesignals are decimated (where decimation is lowpass filtering followed bysample rate compression) such that the final output signals aregenerated at a relatively low sampling rate (e.g., 62.5 Hz) whichprovides increased resolution at the output. Thus, bandwidth is tradedfor resolution in the output signal, thus increasing the signal to noiseratio.

[0009] One aspect of the present invention is an apparatus for measuringblood oxygenation in a subject. The apparatus comprises a first signalsource which applies a first input signal during a first time interval.A second signal source applies a second input signal during a secondtime interval. A detector detects a first parametric signal responsiveto the first input signal passing through a portion of the subjecthaving blood therein. The detector also detects a second parametricsignal responsive to the second input signal passing through the portionof the subject. The detector generates a detector output signalresponsive to the first and second parametric signals. A signalprocessor receives the detector output signal. The signal processordemodulates the detector output signal by applying a first demodulationsignal to a signal responsive to the detector output signal to generatea first output signal responsive to the first parametric signal and byapplying a second demodulation signal to the signal responsive to thedetector output signal to generate a second output signal responsive tothe second parametric signal. Each of the first demodulation signal andthe second demodulation signal comprises at least a first componenthaving a first frequency and a first amplitude and a second componenthaving a second frequency and a second amplitude. The second frequencyis a harmonic of the first frequency. The second amplitude is selectedto be related to the first amplitude to minimize crosstalk from thefirst parametric signal to the second output signal and to minimizecrosstalk from the second parametric signal to the first output signal.In one embodiment, the second amplitude is determined by turning off oneof the first and second signal sources and measuring the crosstalkbetween one of the parametric signals and the non-corresponding outputsignal while varying the second amplitude. A second amplitude isselected that minimizes the measured crosstalk.

[0010] Another aspect of the present invention is a method of minimizingcrosstalk between two signals generated by applying a first pulse and asecond pulse to measure a parameter. The first pulse and the secondpulse are applied periodically at a first repetition rate defining aperiod. The first pulse is generated during a first interval in eachperiod, and the second pulse is generated during a second interval ineach period. The second interval is spaced apart from the firstinterval. The first and second pulses produce first and secondparametric signals responsive to the parameter. The first and secondparametric signals are received by a single detector that outputs acomposite signal responsive to the first and second parametric signals.The method comprises the step of applying a first demodulation signal tothe composite signal to generate a first demodulated output signalwherein the first demodulation signal comprises at least a firstcomponent having a first frequency corresponding to the first repetitionrate. The first component has a first amplitude. The first demodulationsignal further comprises a second component having a second frequencythat is a harmonic of the first frequency. The second component has asecond amplitude which has a selected proportional relationship to thefirst amplitude. The method further includes the step of applying asecond demodulation signal to the composite signal to generate a seconddemodulated output signal. The second demodulation signal comprises thefirst component at the first frequency and the first amplitude andfurther comprises the second component at the second frequency and thesecond amplitude. At least one of the first and second components of thesecond demodulation signal has a selected phase difference with respectto the corresponding one of the first and second components of the firstdemodulation signal. The method further includes the steps of lowpassfiltering the first demodulated output signal to generate a firstrecovered output signal responsive to the first parametric signal; andlowpass filtering the second demodulated output signal to generate asecond recovered output signal responsive to the second parametricsignal.

[0011] Preferably, the selected phase difference is π. Also preferably,the first pulse and the second pulse are generally rectangular pulseshaving a respective duty cycle. The rectangular pulses comprise aplurality of sinusoidal components including a fundamental componentcorresponding to the first frequency and a first harmonic componentcorresponding to the second frequency. The fundamental component has afundamental component amplitude and the first harmonic component has afirst harmonic component amplitude. The first harmonic componentamplitude is related to the fundamental harmonic component amplitude bya first proportionality value. The second amplitude of the secondcomponent of the first demodulation signal is related to the firstamplitude of the first component of the first demodulation signal by asecond proportionality value which is approximately the inverse of thefirst proportionality value.

[0012] The method in accordance with this aspect of the inventionpreferably includes the further steps of sampling the composite signalwhen neither the first pulse nor the second pulse is active to obtain asampled signal; and measuring the sampled signal to determine a noiselevel of the parametric signals.

[0013] In a further embodiment according to this aspect of the presentinvention, the method further includes the steps of performing atransform on the composite signal to generate a spectra of the compositesignal; sampling the spectra at a plurality of frequencies other than atpredetermined ranges of frequencies around the first frequency andaround harmonics of the first frequency; determining an average of themagnitudes of the sampled plurality of frequencies; and comparing theaverage to a selected threshold to determine whether the averagemagnitude exceeds the selected threshold.

[0014] Another aspect of the present invention is a method ofdemodulating a composite signal generated by applying first and secondperiodic pulses of electromagnetic energy to a system having a parameterto be measured and by receiving signals responsive to theelectromagnetic energy after having passed through the system and beingaffected by the parameter being measured. The signals are received as acomposite signal having components responsive to the first and secondpulses. The method comprises the step of applying a first demodulationsignal to the composite signal to generate a first demodulated signal.The first demodulation signal comprises a first component having a firstfrequency corresponding to a repetition frequency of the first andsecond pulses and comprises a second component having a frequency thatis a harmonic of the first frequency. The first component has a firstamplitude and the second component has a second amplitude. The secondamplitude has a predetermined relationship to the first amplitude. Thepredetermined relationship is selected to cause the first demodulatedsignal to have low frequency components responsive only to the firstpulse. The method includes the further step of lowpass filtering thefirst demodulated signal to generate a first output signal. The firstoutput signal varies in response to an effect of the parameter on theelectromagnetic energy received from the first pulse.

[0015] Preferably, the method in accordance with this aspect of theinvention includes the further step of applying a second demodulationsignal to the composite signal to generate a second demodulated signal.The second demodulation signal has first and second componentscorresponding to the first and second components of the firstdemodulation signal. At least one of the first and second components ofthe second demodulation signal has a selected phase relationship withthe corresponding one of the first and second components of the firstdemodulation signal. The method includes the further step of lowpassfiltering the second demodulated signal to generate a second outputsignal. The second output signal varies in response to an effect of theparameter on the electromagnetic energy received from the second pulse.

[0016] Another aspect of the present invention is a pulse oximetrysystem that comprises a modulation signal generator. The modulationsignal generator generates a first modulation signal that comprises afirst pulse that repeats at a first repetition frequency. The firstpulse has a duty cycle of less than 50%. The modulation signal generatorgenerates a second modulation signal comprising a second pulse that alsorepeats at the first repetition frequency. The second pulse has a dutycycle of less than 50%. The second pulse occurs at non-overlapping timeswith respect to the first pulse. Each of the first and second pulsescomprises a plurality of components wherein a first component has afrequency corresponding to the repetition frequency and wherein a secondcomponent has a second frequency corresponding to twice the firstfrequency. The second component has an amplitude which has a firstpredetermined relationship to an amplitude of the first component. Afirst transmitter emits electromagnetic energy at a first wavelength inresponse to the first pulse; and a second transmitter emitselectromagnetic energy at a second wavelength in response to the secondpulse. A detector receives electromagnetic energy at the first andsecond wavelengths after passing through a portion of a subject andgenerates a detector output signal responsive to the receivedelectromagnetic energy. The detector output signal includes a signalcomponent responsive to attenuation of the electromagnetic energy at thefirst wavelength and a signal component responsive to attenuation of theelectromagnetic energy at the second wavelength. A first demodulatormultiplies the detector signal by a first demodulation signal andgenerates a first demodulated output signal. The first demodulationsignal comprises a first component having the first frequency and havinga first amplitude. The first demodulation signal also comprises a secondcomponent having the second frequency and having a second amplitude. Thesecond amplitude has a second predetermined relationship to the firstamplitude. The second predetermined relationship is approximatelyinversely proportional to the first predetermined relationship. A seconddemodulator multiplies the detector signal by a second demodulationsignal and generates a second demodulated output signal. The seconddemodulation signal comprises a first component having the firstfrequency and having the first amplitude. The second demodulation signalfurther comprises a second component having the second frequency andhaving the second amplitude. At least one component of the seconddemodulation signal has a selected phase relationship with acorresponding one component of the first demodulation signal.Preferably, the selected phase relationship is a π phase difference.

[0017] Another embodiment incorporates decimation before demodulation.In yet another embodiment, a multi-channel demodulator, with or withoutpre-demodulation decimation is disclosed.

[0018] In yet another embodiment, an adaptive algorithm is used tocontrol the operation of pre-demodulation decimators andpost-demodulation decimators. The adaptive algorithm may control boththe characteristics of a lowpass filter in the decimator and thedecimation rate provided by a signal rate compressor in the decimator.

[0019] Another embodiment of the invention is a method for selecting asample rate that reduces the interference caused by ambient light.

BRIEF DESCRIPTION OF THE DRAWINGS

[0020] The present invention will be described below in connection withthe accompanying drawing figures in which:

[0021]FIG. 1 illustrates an exemplary block diagram of a representationof a signal processing system in accordance with the present inventionused to determine blood oxygen saturation in a subject;

[0022]FIG. 2 illustrates exemplary waveforms of the current through theLEDs in FIG. 1 and the resulting intensities of the red light and theinfrared light generated by the LEDs;

[0023]FIG. 3 illustrates a block diagram of the overall processingsystem in accordance with the present invention;

[0024]FIG. 4 illustrates a frequency spectra of the first modulationsignal M₁(t) for n=0, 1, 2, . . . , where the horizontal axis representsfrequency and the vertical axis represents the energy in the DC andharmonic components of the signal;

[0025]FIG. 5 illustrates an exemplary spectrum of the first and secondharmonics of the present invention when the fundamental frequency isselected to be 316.7 Hz in comparison to the fundamental and harmonicsof conventional 60 Hz power;

[0026]FIG. 6 illustrates the effect of the value of B on the measuredsignal output Ŝ₂(t) responsive to the red modulation pulses as the valueof B is varied while the infrared modulation pulses are off;

[0027]FIG. 7 illustrates a preferred embodiment of the present inventionimplemented in a digital processing system;

[0028]FIG. 8 illustrates a detailed block diagram of the demodulationportion of the present invention;

[0029]FIG. 9 illustrates a detailed block diagram of the modulationportion of the present invention;

[0030]FIG. 10 illustrates the red drive waveform and the infrared drivewaveform generated by the modulation portion of FIG. 9;

[0031]FIG. 11 illustrates the demodulation waveforms generated by thedemodulation portion of FIG. 8;

[0032]FIG. 12 illustrates a method of time domain sampling the digitaldetection signal during the times when both the red pulses and theinfrared pulses are off to obtain information regarding the level ofambient noise;

[0033]FIG. 13 illustrates a block diagram of a system that performs thetime domain sampling of FIG. 12;

[0034]FIG. 14 illustrates a method of frequency domain sampling todetermine the noise floor at frequencies other than the signalfrequencies;

[0035]FIG. 15 illustrates a block diagram of a system that performs thefrequency domain sampling of FIG. 14;

[0036]FIG. 16 illustrates a block diagram of the overall processingsystem in accordance with a pre-demodulation decimation embodiment ofthe present invention;

[0037]FIG. 17 illustrates a block diagram of a multi-channel processingsystem in accordance with a pre-demodulation decimation embodiment ofthe present invention;

[0038]FIG. 18 illustrates a block diagram of an adaptive multi-channelprocessing system in accordance with a pre-demodulation decimationembodiment of the present invention;

[0039]FIG. 19 illustrates a flowchart of a method for choosing themodulation frequency and decimation rate in order to minimize theaffects of ambient light; and

[0040]FIG. 20 is a graph to be used in connection with graphical methodfor designing a demodulation system to minimize interference due toambient light.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

[0041]FIG. 1 illustrates an exemplary block diagram of a representationof a signal processing system 100 in accordance with the presentinvention used to determine blood oxygen saturation in a subject, suchas, for example, a human subject. In the example presented, themeasurements are performed on a portion of the subject, such as a finger102 illustrated in FIG. 1. An LED modulation circuit 104 drives a pairof back-to-back light emitting diodes (LEDs) 106, 108 by applying aperiodic signal to the two light emitting diodes 106, 108. The LED 106is selected to emit electromagnetic energy in the red visible lightrange, and has a wavelength of, for example, approximately 660nanometers. The LED 108 is selected to emit electromagnetic energy inthe infrared range, and has a wavelength of, for example, approximately905 nanometers. The LED modulation circuit 104 supplies current inalternating directions so that the two LEDs 106, 108 are activated oneat a time. In particular, as illustrated by a current waveform 120 inFIG. 2, current is first applied in a forward direction with respect tothe red LED 106 during a first time interval 122 having a duration τ.Thereafter, no current is applied to either LED during a second timeinterval 124 having a like duration τ. Then, current is applied in aforward direction with respect to the infrared LED 108 during a thirdtime interval 126, also having a duration τ. Then, no current is appliedto either LED during a fourth time interval 128 having a like durationτ. Thereafter, the current is again applied in the forward direction forthe red LED 106 during a fifth time interval 130 which corresponds tothe first time interval 122. It can be seen that the overall cyclerepeats with a period of duration T equal to 4τ. The red LED 106 emitslight only when the current is applied in the forward direction withrespect to the red LED 106. Thus, as illustrated by a red intensitywaveform 132, the red LED 106 emits light as a pulse 134 during thefirst time interval 122 and as a pulse 136 during the fifth timeinterval 130, and so on. The red pulses repeat with a periodicity equalto T. Similarly, the infrared LED 108 emits infrared light only when thecurrent is applied in the forward direction with respect to the infraredLED 108. Thus, as illustrated by an infrared intensity waveform 140, theinfrared LED 108 emits infrared light as a pulse 142 during the thirdinterval 126. A next infrared pulse 144 occurs at an interval T afterthe infrared pulse 142. Thus, the infrared pulses also repeat with aperiodicity equal to T. It can be seen that the red pulses and theinfrared pulses each have a duty cycle of 25%, and the red pulses andthe infrared pulses are separated by intervals of one-fourth of eachperiod T (i.e., the beginning of one pulse occurs an interval τ afterthe end of the previous pulse).

[0042] As further illustrated in FIG. 1, the electromagnetic energypulses from the red LED 106 and the infrared LED 108 are applied to thefinger 102. A detector 150 is positioned to receive the electromagneticenergy after the energy has passed through a portion of the finger 102.The detector 150 is selected to be responsive to both the red light andthe infrared light and to generate an output signal responsive to theintensity of the energy received from each source. An exemplary currentoutput signal from the detector 150 is represented by a waveform 152 inFIG. 2. As illustrated, the detector signal waveform 152 comprises afirst pulse 154 responsive to the first red pulse 134, a second pulse156 responsive to the infrared pulse 142 and a third pulse 158responsive to the second red pulse 136. During the time between thefirst pulse 154 and the second pulse 156, the detector signal waveform152 comprises noise 160, and during the time between the second pulse156 and the third pulse 158, the detector signal waveform 150 comprisesnoise 162. The signal pulses 154, 156 and 158 also include noisesuperimposed thereon. Although shown as repeating noise, it should beunderstood that the noise varies with time. For example, noise caused byambient light will vary with a periodicity corresponding to the 50 Hz or60 Hz power frequency and their harmonics, particularly when the ambientlight is provided by fluorescent lights which generate significant noiseat the first harmonic (i.e., 100 Hz or 120 Hz) and the third harmonic(i.e., 200 Hz or 240 Hz).

[0043] The output of the detector 150 is applied as an input to a signalprocessor block 170 which processes the detector signal and generates afirst signal Ŝ₁(t) responsive to the detected intensity of the red lightincident on the detector 150 and generates a second signal Ŝ₂(t)responsive to the detected intensity of the infrared light incident onthe detector 150. As illustrated, the signal processing block 170 issynchronized with the LED modulator 104 via a set of control lines 180.As will be discussed below, the control lines 180 advantageouslycommunicate signals which provide timing information that determineswhen to activate the red LED 106 and when to activate the infrared LED108.

[0044]FIG. 3 is a pictorial representation of a model of an exemplarysystem which incorporates the present invention. The red LED 106provides a light intensity represented as I_(RD), and the infrared LED108 provides a light intensity represented as I_(IR). The effects ofturning the LEDs 106, 108 on and off on periodic bases are modeled by afirst multiplier or modulator 190 which applies a first modulationsignal M₁(t) to the red light intensity to generate a modulated redsignal I_(RDMOD)(t) and by a second multiplier or modulator 192 whichapplies a second modulation signal M₂(t) to the infrared light intensityto generate a modulated infrared signal I_(IRMOD)(t). The modulatedlight red signal and the modulated infrared signal are applied to thefinger 102, or other body portion, as described above. The finger 102has blood flowing therein and is represented in FIG. 3 as a block 102.The blood in the finger 102 has a volume and scattering components whichvary throughout each cardiac cycle. The blood carries oxygen and othermaterials therein. The oxygen content is a function of both the bloodvolume and the concentration of the oxygen in the blood volume. Theconcentration of the oxygen in the blood volume is generally measured asblood oxygen saturation for reasons which are described in full in theabove-identified issued U.S. Pat. Nos. 5,482,036 and 5,490,505. Asfurther described in the two referenced patents, the blood oxygensaturation is determined by comparing the relative absorption of the redlight and the infrared light in the finger 102. The comparison iscomplicated by the noise caused by movement, ambient light, lightscattering, and other factors.

[0045] In FIG. 3, a pair of signals S₁(t) and S₂(t) represent the effectof the time-varying volume and scattering components of the blood in thefinger 102 on the red light and the infrared light, respectively,passing through the finger 102 from the LEDs 106, 108 to the detector150. The red light signal portion S₁(t) is caused by the variableattenuation of the red light passing through the finger 102. Theinfrared light signal portion S₂(t) is caused by the variableattenuation of the infrared light passing through the finger 102. Toshow the effect of the variable attenuations, the signal portion S₁(t)is illustrated as being applied to a first attenuation modulator 191which multiplies the signal S₁(t) by the modulated red outputI_(RDMOD)(t) of the fist modulator 190. Similarly, the infrared lightsignal portion S₂(t) is illustrated as being applied to a secondattenuation modulator 193 which multiplies the signal S₂(t) by themodulated infrared output I_(IRMOD)(t) of the second modulator 192. Theoutputs of the first and second attenuation modulators 191, 193 areprovided to the receiving photodetector 150. The photodetector 150 ismodeled as adder 194 and an adder 196. The outputs of the first andsecond attenuation modulators 191, 193 are provided to the adder 194 togenerate a composite signal M(t) where:

M(t)=S ₁(t)M ₁(t)+S ₂(t)M ₂(t).  (1)

[0046] The signal M(t) from the adder 194 is provided to the adder 196where the signal M(t) is added to a signal n(t) which represents acomposite noise signal caused by ambient light, electromagnetic pickup,and the like, which are also detected by the photodetector 150. Theoutput of the adder 196 is a signal M′(t)=M(t)+n(t) which includes noisecomponents as well as the signal components. The noise componentsinclude DC components and harmonics of the power line frequency thatappear in the ambient light. In addition, as will be discussed in moredetail below, the signal M′(t) may also include noise at higherfrequencies caused, for example, by other devices such aselectrocauterization equipment, or the like.

[0047] The M′(t) signal output of the third adder 196 (i.e., the outputof the detector 150) is applied to the input of the signal processingblock 170. Within the signal processing block 170, the signal M′(t) isfirst passed through a fixed gain amplifier 197 and then through ananalog bandpass filter 198. The analog bandpass filter 198 has apassband selected to pass signals in the range of 20 Hz to 10,000 Hz.Thus, the analog bandpass filter 198 removes a significant portion ofthe noise below 10 Hz. The signal components responsive to the bloodoxygen saturation are frequency shifted by the operation of the twomodulation signals M₁(t) and M₂(t) and are passed by the analog bandpassfilter 198.

[0048] In the preferred embodiment, the output of the analog bandpassfilter 198 is sampled by an analog-to-digital converter 199 andconverted therein to digital signals. For example, the signals arepreferably sampled at 46,875 samples per second. The output of theanalog-to-digital converter 199 is a signal MF(k).

[0049] The signal MF(k) is provided as a first input to a firstdemodulating multiplier 210. The signal MF(k) is also provided as afirst input to a second demodulating multiplier 212. A firstdemodulating signal D₁(k) is provided as a second input to the firstdemodulating multiplier 210, and a second demodulating signal D₂(k) isprovided as a second input to the second demodulating multiplier 212.The output of the first demodulating multiplier 210 is provided as aninput to a first lowpass filter 220, and the output of the seconddemodulating multiplier is provided as an input to a second lowpassfilter 222. The bandwidths of the lowpass filters 220, 222 arepreferably approximately 10 Hz.

[0050] The output of the first lowpass filter 220 is a signal Ŝ₁(t),which, as discussed below, is an estimate of the signal S₁(t). Theoutput of the second lowpass filter 222 is a signal Ŝ₂(t), which, asdiscussed below, is an estimate of the signal S₂(t). As will be shownbelow, the selection of the first demodulating signal D₁(k) and thesecond demodulating signal D₂(k) in accordance with the presentinvention substantially reduces or eliminates the effects of noise inthe two output signals Ŝ₁(t) and Ŝ₂(t) and also substantially reduces oreliminates crosstalk between the two signals.

[0051] In the preferred embodiment of the present invention, the samplerates of the outputs of the lowpass filter 220 and the lowpass filter222 are compressed by respective sample rate compressors 221 and 223. Inparticular, the sample rate compressors 221, 223 reduce the sample rateby 750 to a sample rate of, for example, 62.5 Hz to provide an outputwhich can be further processed in accordance with the methods andapparatuses described in the above-referenced patents. The sample ratecompressions which occur in the sample rate compressors 221, 223 reducethe rate at which the output signals Ŝ₁(t) and Ŝ₂(t) need to beprocessed while maintaining the sample rate well above the 0-10 Hzfrequency content of the signals of interest. The outputs of the filters220, 222, or the sample rate compressors 221, 223, if included, areprovided on respective output lines 224 and 226.

[0052] In order to facilitate an understanding of how the presentinvention operates in demodulating the output signal MF(k) from theanalog-to-digital converter 199, the modulation signals M₁(t) and M₂(t)will first be described in terms of their frequency components. Oneskilled in the art will appreciate that the modulation signals M₁(t) andM₂(t) can each be represented as a Fourier cosine series expansion(e.g.,${\sum\limits_{n = 0}^{\infty}{a_{n}\quad {\cos ( {n\quad \omega \quad t} )}}},$

[0053] where ω=2π/T) representing the fundamental and harmonicfrequencies of the rectangular signal pulses. One skilled in the artwill understand that the Fourier series expansion includes phases;however, by suitably selecting the time origin, the phases are set tozero. A component which is 180° out of phase with a correspondingcomponent will advantageously be represented by a minus sign before thecoefficient.

[0054]FIG. 4 illustrates a frequency spectra of the first modulationsignal M₁(t) for n 0, 1, 2, . . . , where the horizontal axis representsfrequency, with the energy in the DC component along the vertical axisand increasing harmonics of the fundamental frequency along thehorizontal axis. The length of each component of M₁(t) along thevertical axis represents the energy E(n) in each component of thefrequency spectra. The first component to the right of the vertical axisis at the fundamental frequency (i.e., 1/T), which is designated hereinas f₀; however, it should be understood that the fundamental frequencyf₀ corresponds to n=1. The second component to the right of the verticalaxis is the first harmonic f₁ (i.e., n=2), which has a frequency whichis twice the fundamental frequency. The third component to the right ofthe vertical axis is the second harmonic f₂ (i.e., n=3), which has afrequency which is three times the fundamental frequency. The componentsto the right of the second harmonic are numbered accordingly. (Note,other conventions identify the fundamental frequency as the firstharmonic, and designate the second harmonic as the frequency that istwice the fundamental frequency. The identification of the fundamentalfrequency as f₀ is used in the discussion that follows.)

[0055] In FIG. 4, a modulation envelope 230 is shown in dashed lines.The modulation envelope 230 represents the magnitudes of the fundamentaland the harmonics of the signal M₁(t). The shape of the envelope isdetermined by the modulation signal M₁(t) which, for a repeatingrectangular pulse train starting at time t=0 and having a normalizeamplitude of 1, can be expressed as: $\begin{matrix}{{M_{1}(t)} = {\frac{\tau}{T}\quad {\sum\limits_{n = 0}^{\infty}{\sin \quad {c( \frac{n\quad \tau}{T} )}\quad {\cos ( \frac{2\quad \pi \quad {nt}}{T} )}}}}} & (2)\end{matrix}$

[0056] Where sinc is the function (sin πx)/πx (i.e.,sinc(nτ/T)=sin(nπτ/T)/(nπτ/T)). In the example shown, τ=¼T. (Note thatfor sampled signals, the envelope is more accurately represented as sinα/sin β; however, as well known in the art, for the frequencies ofinterest, the sinc function is a suitable approximation.) Thus, thefrequency spectra has nulls at n=4, n=8, n=12, and so on, correspondingto the third harmonic f₃, the seventh harmonic f₇, the eleventh harmonicf₁₁, and so on. Note that Equation 2 is an idealized form of theequation for M₁(t), and that in general: $\begin{matrix}{{M_{1}(t)} = {\frac{\tau}{T}{\sum\limits_{n = 0}^{\infty}{a_{n}\sin \quad c\quad \frac{n\quad \tau}{T}\quad ^{{- j}\quad \omega_{0}n\quad t}}}}} & (3)\end{matrix}$

[0057] where a_(n) is a complex number. In the discussion that follows,the values of a_(n) are assumed to be real numbers only.

[0058] A similar frequency spectra (not shown) for the modulation signalM₂(t) is determined by the expression: $\begin{matrix}{{M_{2}(t)} = {\frac{\tau}{T}\quad {\sum\limits_{n = 0}^{\infty}{( {- 1} )^{n}\quad \sin \quad {c( \frac{n\quad \tau}{T} )}\quad {\cos ( \frac{2\quad \pi \quad n\quad t}{T} )}}}}} & (4)\end{matrix}$

[0059] An envelope for the frequency spectra of second modulation signalM₂(t) will have the same magnitudes; however, it should be understoodthat because of the (−1)^(n) term in the expression for M₂(t), thefundamental f₀ and every even harmonic (i.e., f₂, f₄, etc.) are 180° outof phase with the corresponding harmonic of the first modulation signalM₁(t).

[0060] In FIG. 3, the analog-to-digital converter 199 converts thesignal M′(t) to a sequence of sampled digital values MF(k) at a samplingrate of, for example, 46,875 samples per second. As discussed above, thefirst demodulating multiplier 210 multiplies the output MF(k) of theconverter 199 by the first demodulating signal D₁(k) to generate thefirst output sequence Ŝ₁(k), and the second demodulating multiplier 212multiplies the output MF(k) by the second demodulating signal D₂(k) togenerate the second output sequence Ŝ₂(k). The multiplication by themultipliers 210, 212 can also be expressed as follows:

Ŝ ₁(k)=LP[MF(k)D ₁(k)]  (5)

[0061] and

Ŝ ₂(k)=LP[MF(k)D ₂(k)]  (6)

[0062] where LP is the transfer function of the lowpass filter 220 andof the lowpass filter 222. If, for simplicity, the noise is assumed tobe zero, then:

M′(t)=S ₁(t)M ₁(t)+S ₂(t)M ₂(t)  (7)

[0063] Therefore:

Ŝ ₁(k)=LP[[S ₁(k)M ₁(k)+S ₂(k)M ₂(k)]D ₁(k)]  (8)

[0064] and thus

Ŝ ₁(k)=LP[[S ₁(k)M ₁(k)]D ₁(k)+[S ₂(k)M ₂(k)]D ₁(k)]  (9)

[0065] Similarly:

Ŝ ₂(k)=LP[[S ₂(k)M ₂(k)]D ₁(k)+[S ₁(k)M ₁(k)]D ₂(k)]  (10)

[0066] Since LP is a linear operator, the right-hand side of Equations 9and 10 can be split into two terms. The first term on the right-handside of each of Equations 9 and 10 above is the desired signal portionof the equation, and the second term on the right-hand side of each ofthe equations is the crosstalk portion. Thus, in order to reduce thecrosstalk to zero, the second term of each of Equations 9 and 10 is setto zero:

LP[S ₂(k)M ₂(k)D ₁(k)]=0  (11)

[0067] and

LP[S ₁(k)M ₁(k)D ₂(k)]=0  (12)

[0068] By setting the second terms to zero, Equations 9 and 10 reduceto:

Ŝ ₁(k)=LP[S ₁(k)M ₁(k)D ₁(k)]  (13)

[0069] and

Ŝ ₂(k)=LP[S ₂(k)M ₂(k)D ₂(k)]  (14)

[0070] One goal of the present invention is to select the demodulatingsignals D₁(k) and D₂(k) to satisfy Equations 11 and 12 to thereby reduceEquations 9 and 10 to Equations 13 and 14. This is accomplished byutilizing Equations 2 and 3 to simplify the two equations by selectivelyusing components of the two modulating signals M₁(t) and M₂(t) togenerate the demodulating sequences D₁(k) and D₂(k).

[0071] In order to simplify the discussion, Equation 2 can be rewrittenas: $\begin{matrix}{{M_{1}(t)} = {\sum\limits_{n = 1}^{\infty}{{E(n)}\quad {\cos ( {n\quad \omega \quad t} )}}}} & (15)\end{matrix}$

[0072] where E(n) is the sinc envelope for the fundamental frequency f₀(n=1) and the harmonics f₁ (n=2), f₂ (n=3), and so on, where cos(nωt)represents the cosine term cos(2πnt/T), where ω=2π/T. (Note, asdiscussed above, for discrete sampled signals, the actual envelope ofE(n) is a sin α/sin β function; however, for the frequencies ofinterest, the sinc function is a suitable representation.)

[0073] As discussed above, the DC term (n=0) does not need to beconsidered because of the operation of the filter 198, and theanalog-to-digital converter 199, as well as the action of thedemodulation, which shift any unwanted DC or low frequency signalshaving a frequency less than approximately 10 Hz (hereinafter near-DCsignals) to higher frequencies before lowpass filtering. As a furthersimplification, the magnitude of the fundamental term in Equation 15 isnormalized to a value of 1 (i.e., E(1)=1). Note that the normalizationresults in the need for a scale factor, which will be discussed below.

[0074] Thus, Equation 15 becomes:

M ₁(t)=cos ωt+a cos 2ωt+b cos 3ωt+c cos 4ωt+ . . .  (16)

[0075] The demodulation signal D₁(t) is defined as:

D ₁(t)=cos ωt+B cos 2ωt  (17)

[0076] For reasons set forth below, only the first two cosine terms areneeded.

[0077] Similarly, the second modulating signal M₂(t) becomes:

M ₂(t)=−cos ωt+a cos 2ωt−b cos 3ωt+c cos 4ωt+ . . .  (18)

[0078] and the second demodulating signal D₂(t) is defined as:

D ₂(t)=−cos ωt+B cos 2ωt  (19)

[0079] Note that the signs of the fundamental and odd harmonics inEquation 18 are 180° out of phase with the corresponding terms inEquation 16.

[0080] Note, as will be developed more fully below, by including onlythe fundamental (cos ωt) and the first harmonic (cos 2ωt) in each of thedemodulation signals, only the signals proximate to the fundamental andfirst harmonic need to be considered. By eliminating higher harmonics,the effects of the higher harmonics of the power line frequency are alsoeliminated in the output signals generated by the present invention.

[0081] Assume that the filter 198 and the analog-to-digital converter199 do not affect the magnitude of the signal MF(k) with respect toM′(t) for the frequencies having significant energy. Therefore, startingwith Equation 7 above, M′(t) can be written as:

M′(t)=S ₁(t)[cos ωt+a cos 2ωt+b cos 3ωt+ . . . ]+S ₂(t)[−cos ωt+a cos2ωt−b cos 3ωt+ . . . ]  (20)

[0082] When the first demodulating multiplier 210 multiplies M′(t) byD₁(t), the terms on the right-hand side of Equation 20 are multiplied bythe terms on the right-hand side of Equation 17. Thus:

M′(t)D ₁(t)=S ₁(t)[cos ωt+a cos 2ωt+b cos 3ωt+ . . . ][cos ωt+B cos2ωt]+S ₂(t)[−cos ωt+a cos 2ωt−b cos 3ωt+ . . . ][cos ωt+B cos 2ωt]  (21)

[0083] The term S₁(t)[cos ωt+a cos 2ωt+b cos 3ωt+ . . . ][cos ωt+B cos2ωt] is the signal term which is to be preserved, and the termS₂(t)[−cos ωt+a cos 2ωt−b cos 3ωt+ . . . ][cos ωt+B cos 2ωt] is thecrosstalk term to be eliminated.

[0084] Expanding the crosstalk term from Equation 21, generates:

crosstalk=S ₂(t)[−cos 2ωt−B cos ωt cos 2ωt+a cos 2ωt cos ωt+aB cos 2ωt−bcos 3ωt cos ωt−bB cos 3ωt cos 2ωt+ . . . ]  (22)

[0085] Using the identity, cos(x)cos(y)=½[cos(x+y)+cos(x−y)], thecrosstalk term from Equation 22 becomes:

crosstalk=S ₂(t)[−½(cos 2ωt+1)+((a−B)/2)[cos 3ωt+cos ωt]+(aB/2)[cos4ωt+1]−(b/2)[cos 4ωt+cos 2ωt]−(bB/2)[cos 5ωt+cos ωt]+ . . . ]  (23)

[0086] The remaining terms in Equation 23 will all have a factor of cosωt or higher. Thus, Equation 23, when fully expanded only includesnear-DC terms:

crosstalk_(DC) =LP[S ₂(t)[(aB/2)−½]]  (24)

[0087] where S₂(t) corresponds to the infrared portion of the originalplethysmograph signal which has a bandwidth of interest of approximately0 to 10 Hz. Any components present above 10 Hz will be eliminated by theaction of the lowpass filter 220. Thus, it can be seen that only thesignals of interest are folded back to DC or near-DC. By using thelowpass filter 220, the DC terms and near-DC terms can be isolated sothat only the DC terms and near-DC terms of the crosstalk are presentedat the output of the lowpass filter 220. Thus, in order to eliminate thecrosstalk, the crosstalk terms in Equation 24 need to be set to zero:

LP[S ₂(t)[aB/2−½]]=0  (25)

[0088] Thus:

B=1/a  (26)

[0089] The result in Equation 26 can also be expressed using a geometricinterpretation of vector projection (i.e., dot products) of S₂(t) andS₁(t) wherein the projection of S₂(t) onto D₁(t) is equal to zero andthe projection of S₂(t) onto D₂(t) is maximized. In other words, expressS₁(t), S₂(t), D₁(t) and D₂(t) as vectors of samples in an n-dimensionalsample space (e.g., S₁(t) is represented as a vector S₁ of samplesS₁(k)). For example, in a preferred embodiment, n=148, and thus S₁, S₂,D₁ and D₂ are vectors of 148 samples each. The first crosstalk term isS₁ D₂. The second crosstalk term is S₂·D₁. The first signal output isS₁·D₁. The second signal output is S₂·D₂. Select the vectors D₁ and D₂to drive the crosstalk terms to zero.

[0090] The relationship in Equation 26 also works to preserve the signalterm. In particular, the signal term in Equation 21 can be expanded andlowpass filtered in the same manner as the crosstalk term to obtain:

signal=Ŝ ₁(t)=LP[S ₁(t)[(aB/2)+½]]  (27)

[0091] Using the relationship from Equation 26, then Equation 27becomes:

signal=Ŝ ₁(t)=LP[S ₁(t)[(a/2a)+½]=LP[S ₁(t)]=S ₁(t)  (28)

[0092] It can be readily shown that the same relationship holds for thecrosstalk term and the signal term for the signal S₂(t) by defining thesecond demodulation signal D₂(t) as:

D ₂(t)=−cos ωt+B cos 2ωt  (29)

[0093] and multiplying M₂(t) by D₂(t). After expanding the crosstalk andsignal terms and eliminating the terms above 10 Hz, it can be shown thatby selecting B=1/a, the crosstalk term is canceled and the signal termS₂(t) is recovered.

[0094] From the foregoing, it can be seen that by choosing therelationship between the magnitude of B as the reciprocal of a, then thecrosstalk terms are eliminated and the signal terms are preserved. Notethat neither A nor B is an absolute value. As set forth in Equation 16,a is the magnitude of the cos 2ωt term of M₁(t) when the magnitude ofthe cos ωt term of M₁(t) is normalized to 1. Similarly, from Equation17, B is the magnitude of the cos 2ωt term of D₁(t) when the cos ωt termof D₁(t) is normalized to 1.

[0095] It should be understood that both D₁(t) and D₂(t) can includehigher harmonic terms; however, such additional terms could result inincreased sensitivity to the noise of fluorescent lights and the likebecause of the harmonics of the 60 Hz power line frequency (or the 50 Hzpower line frequency in other countries). For example, FIG. 5illustrates an exemplary spectrum of the first and second harmonics ofthe present invention when the fundamental frequency is selected to be316.7 Hz. Thus, the first harmonic frequency is 633.4 Hz. Note that thevariations in the signals caused by blood flow throughout a cardiaccycle causes the fundamental and harmonics of modulation frequency to besurrounded by sidebands representing the frequency content of theplethysmograph. For example, in FIG. 5, the first and second harmonicsare at 316.7 Hz and 633.4 Hz, ±10 Hz.

[0096] As further illustrated in FIG. 5, the conventional 60 Hz powerline frequency has harmonics at 120 Hz, 180 Hz, 240, etc. Thus, thenearest harmonics of the power line frequency to the first harmonic ofthe present invention are at 300 Hz and 360 Hz, and the nearestharmonics of the power line frequency to the second harmonic of thepresent invention are 600 Hz and 660 Hz. Similarly, if used in a countryhaving a 50 Hz power line frequency, the nearest harmonics to the firstharmonic of the present invention are 300 Hz and 350 Hz, and the nearestharmonics to the second harmonic of the present invention are 600 Hz and650 Hz. Even if the power frequency were to vary by up to 1.5 percent,the noise generated by the ambient light from fluorescent lamps, or thelike, would not be at the first and second harmonic frequencies of thepresent invention. The fundamental frequency has thus been selected toavoid power line caused ambient noise at the first and second harmonicfrequencies.

[0097] The foregoing discussion assumed that the filter 198 did notsignificantly affect the amplitude of the filtered signal. If the filter198 does have an affect on the amplitude, then B will be a constanttimes the value of B determined above:

B=k/a  (30)

[0098] where k depends on the relative attenuation of the first harmonicand the second harmonic through the filter 198.

[0099] Although the value of the coefficient B can be calculated as setforth above, the calculations may be complicated if the filter 198 orthe modulators 190, 192 introduce phase changes which cause thecalculations to be performed on complex numbers. For example, if themodulation signals M₁(t) and M₂(t) are not rectangular waves which have25% duty cycles and which are precisely 180° out of phase, asillustrated herein, then the coefficients of the frequency components ofthe modulation signals may be complex to account for the phaserelationships, and thus, the coefficients of the demodulation signalsmay be complex.

[0100] As illustrated in FIG. 6, the value of B can also be determinedempirically by performing a initial measurement with one channel (i.e.,either the red pulse or the infrared pulse turned off) and minimizingthe crosstalk. In particular, during the initial measurement, thewaveform 140 in FIG. 2 is set to a continuous zero value so that noinfrared pulses are generated. Thus, the detector 150 (FIG. 1) receivesonly the light generated by the red LED 106. Thus, M₂(t) is set to zero,and Equation 10 for Ŝ₂(t) becomes:

Ŝ ₂(t)=LP[S ₁(t)M ₁(t)D ₂(t)]  (31)

[0101] It can be seen that Ŝ₂(t) includes only a crosstalk portion,which can be measured on the output from the second lowpass filter 222.Thus, by varying the value B while monitoring the magnitude or the RMS(root-mean-squared) value of the output signal Ŝ₂(t), a minimummagnitude Ŝ₂(t)_(min) for the output signal Ŝ₂(t) can be found whichcorresponds to the best value B_(BEST) for B. In an ideal system, thebest value for B corresponds to a zero value for the output signalŜ₂(t); however, in a real environment, the best value of B maycorrespond to a non-zero value for Ŝ₂(t) (i.e., a minimum error forŜ₂(t)). It should be understood that the value of B_(BEST) can also bedetermined by turning off the red LED 106 and varying B while monitoringŜ₁(t) until Ŝ₁(t) is minimized.

[0102] From the foregoing, it can be seen that the effect of themodulation signals D₁(t) and D₂(t) is to shift the DC or near-DC noiseterms up in frequency while shifting the signals of interest at theharmonics back to DC or near-DC, which in effect interchanges the noisespectra and the signal spectra so that the noise spectra can beeliminated by the action of the lowpass filters 220, 222, leaving onlythe signals of interest.

[0103]FIG. 7 illustrates a preferred embodiment of the present inventionwhich implements the functions described above in a digital system.Preferably, the digital system comprises a digital signal processor (notshown), and the blocks described herein comprise data structures withinthe digital signal processor and software routines that implement theprocesses described below. In particular, the present inventioncomprises an LED demodulation block 300 which receives a digitalconfiguration signal on a bus 310, a clock signal on a line 312 and adigital detector signal on a bus 314 as inputs. The digitalconfiguration signal bus 310 provides a way to change the configurationof the LED demodulation block 300 to accommodate different LEDs anddifferent detection algorithms. Preferably, the clock signal on the line312 is a 46,875 Hz (46.875 kHz) square wave signal which is used tosynchronize the timing functions of the present invention. The digitaldetector signal on the line 314 is the output of the analog-to-digitalconverter 199. The analog-to-digital converter 199 is connected to theoutput of the detector 150 (via the amplifier 197 and the filter 198)and samples the output of the detector 150 at 46,875 samples per secondto provide a stream of sampled digital values of the red light andinfrared light incident on the detector 150.

[0104] The LED modulation block 300 generates a demodulated red signaloutput on a bus 340 and generates a demodulated infrared signal outputon a bus 342. The demodulated red signal output is passed through thelow pass filter 220 and is output therefrom as the signal Ŝ₁(t). Thedemodulated infrared signal output is passed through the low pass filter222 and is output therefrom as the signal Ŝ₂(t). As further illustratedin FIG. 8, the LED demodulation block 300 comprises a modulo-M block350, an LED demodulation state table block 352, the first demodulatingmultiplier 210 and the second demodulating multiplier 212.

[0105] The modulo-M block 350 receives the main 46,875 Hz clock signalon the line 312 as one input and receives a MODULUS signal on a bus 354as a second input. The bus 354 forms a portion of the configuration bus310. The modulo-M block 350 divides the clock signal by the MODULUSsignal and generates a RESIDUE signal (described below) on a bus 356which is provided as one input to the LED modulation state table block352. The LED modulation state table block 352 also receives theconfiguration signals on the configuration bus 310.

[0106] The LED demodulation state table is responsive to the residuesignal and the configuration signals to generate the first demodulatingsignal D₁(t) on a bus 360 and to generate the second demodulating signalD₂(t) on a bus 362. The first demodulating signal D₁(t) is provided asone input to the first demodulating multiplier 210, as described above.The second demodulating signal D₂(t) is provided as one input to thesecond demodulating multiplier 212, as described above. The firstdemodulating multiplier 210 and the second demodulating multiplier 212receive the digital detector signal on the line 314 as respective secondinputs. The demodulating multipliers 210, 212 multiply the digitaldetector signal by the first demodulating signal D₁(t) and the seconddemodulating signal D₂(t), respectively, to generate a demodulated redsignal and a demodulated infrared signal on the buses 340 and 342,respectively. Because the outputs of the two demodulating multipliers210 and 212 include the terms cos ωt, cos 2ωt, and higher, thedemodulated signals on the buses 340 and 342 are provided as respectiveinputs to the low pass filters 220 and 222 to pass only the near-DCterms, as discussed above. The outputs of the lowpass filters 220 and222 on the buses 344 and 346, respectively, are the Ŝ₁(t) signal and theŜ₂(t) signal which contain only the near-DC terms, which, in accordancewith the discussion presented above represent the original input signalsS₁(t) and S₂(t) with the unwanted noise substantially reduced oreliminated. The two signals Ŝ₁(t) and Ŝ₂ (t) are then applied tocomputation circuitry (not shown) which computes the blood oxygensaturation and other cardiographic parameters in a manner described inthe above-cited U.S. Pat. Nos. 5,482,036 and 5,490,505.

[0107] The residue signal generated as the output from the modulo-Mblock 350 is a multiple bit signal that counts from 0 to MODULUS-1. Inthe preferred embodiment described herein, MODULUS has a value of 148.Thus, the RESIDUE output of the modulo-M block 350 counts from 0 to 147.The RESIDUE output of the modulo-M block 350 is a number that isprovided as the input to the LED demodulation state table block 352. Asillustrated in FIG. 9, the RESIDUE output on the bus 356 corresponds tothe signal 180 in FIG. 1 and is also provided to the input of an LEDmodulation state table block 370 which, together with an LED drivercircuit 372, comprise the modulation block 104 (FIG. 1) which generatesthe drive signals to the red LED 106 and the infrared LED 108. Asdescribed above, the red LED 106 and the infrared LED 108 generate themodulation signals M₁(t) and M₂(t), respectively, which effectivelyoperate as carriers for the plethysmograph waveform to be measured. Inparticular, as illustrated by a red drive timing waveform 374 and by ainfrared drive timing waveform 376 in FIG. 10, the modulation statetable block 370 generates a red signal pulse 378 during the time whenthe RESIDUE signal increments from 0 to 36. Then, the modulation statetable block 370 generates neither a red signal pulse nor an infraredsignal pulse during the time when the RESIDUE signal increments from 37to 73. Then, the modulation state table block 370 generates the infraredsignal pulse 380 during the time when the RESIDUE signal increments from74 to 110. Then, the modulation state table block 370 again generatesneither a red signal pulse nor an infrared signal pulse during the timewhen the RESIDUE signal increments from 111 to 147. The RESIDUE signalthen resets to 0 and the process repeats continuously.

[0108] The red signal pulse 378 and the infrared signal pulse 380 fromthe modulation state table block 370 are provided as inputs to the LEDdriver circuit 372 which turns on the red LED 106 when the red signalpulse 376 is active and turns on the infrared LED 108 when the infraredsignal pulse 378 is active by generating the current waveform 120illustrated in FIG. 2. The circuitry for converting the red signal pulse376 and the infrared signal pulse 378 to the bi-directional currentpulses of the waveform 120 is conventional and does not need to bedescribed herein.

[0109] In the preferred embodiment, the LED demodulation state tableblock 352 implements demodulation equations which generally correspondto the Equations 17 and 19 described above. In particular, the LEDdemodulation state table block 352 receives the RESIDUE as one input tothe state table and steps through the state table based upon the currentvalue of the RESIDUE. The LED demodulation state table block 352generates two output values for each value of the RESIDUE, wherein thefirst output value is the first demodulation signal D₁(t) on the signalbus 360, and the second output value is the second demodulation signalD₂(t) on the signal bus 362.

[0110] In particular, the LED demodulation state table block 352implements the following forms of the demodulation signal D₁(t) and theD₂(t) equations: $\begin{matrix}\begin{matrix}{{D_{1}(t)} = {- {{SCL}( {{\cos \lbrack {2\quad \pi \quad {t( \frac{R - 18.5 - {{HW}\quad \Delta}}{Modulus} )}} \rbrack} +} }}} \\ {{HWD}( {\cos \lbrack {4\quad \pi \quad {t( \frac{R - 18.5 - {{HW}\quad \Delta}}{Modulus} )}} \rbrack} )} )\end{matrix} & (32) \\{and} & \quad \\\begin{matrix}{{D_{2}(t)} = {- {{SCL}( {{\cos \lbrack {2\quad \pi \quad {t( \frac{R - 18.5 - {{HW}\quad \Delta}}{Modulus} )}} \rbrack} -} }}} \\ {{HWD}( {\cos \lbrack {4\quad \pi \quad {t( \frac{R - 18.5 - {{HW}\quad \Delta}}{Modulus} )}} \rbrack} )} )\end{matrix} & (33)\end{matrix}$

[0111] In Equations 32 and 33, the value SCL is a scale factor whichdetermines the magnitudes of the two demodulation signals and which isused to compensate for the normalization discussed above and tocompensate for other factors, such as, for example, non-idealrectangular pulses. The method of determining the scale factor will beset forth below. In one particularly preferred embodiment, the value ofSCL is 2.221441469. The value HWD is a hardware distortion factor, whichcorresponds to the value of B discussed above. The determination of thevalue B was described above, and will be described again below inconnection with this preferred embodiment. In one particularly preferredembodiment where the pulses applied to the red LED 106 and the infraredLED 108 are idealized rectangular waves having 25% duty cycles, thevalue of HWD can be calculated to be 1.414213562. This ideal value forHWD can be determined by recognizing that the value of the coefficient Afor the cos 2ωt terms in Equations 16 and 18 is determined by the sincfunction. When the coefficient of the cos ωt term is normalized to 1, asin the two equations, then the value of the coefficient a is equal to{square root}{square root over (2)}/2. Thus, the ideal value for B(i.e., HWD) is {square root}{square root over (2)}. Of course, theactual value of the coefficient B, and thus HWD, will vary when the redpulses and the infrared pulses are not true rectangular waves. Since, inactual embodiments, the pulses will have finite rise times and falltimes, the optimum value of HWD is preferably found empirically in themanner described below.

[0112] The value 18.5 in Equations 32 and 33 is used to align thedemodulation waveforms with the modulation waveforms so that the peak ofthe cosine functions corresponds to the midpoints of each of themodulation waveforms. The value HWΔ is a hardware delay factor which maybe needed in certain embodiments to compensate for delays in the analogprocessing, the digital processing or both, which cause the demodulationsignals D₁(t) and D₂(t) to be out of phase with the modulation signalsM₁(t) and M₂(t). In an ideal environment, the value of the hardwaredelay factor is 0. However, in one particularly preferred embodiment,the value of the hardware delay factor is 39. The modulus was describedabove and is basically the number of steps in each period of thewaveforms. In the embodiment described herein, the modulus is 148. Thevalue R is the RESIDUE, which varies from 0 to modulus-1, and thus, inthe preferred embodiment, R varies from 0 to 147.

[0113] In operation, the clock signal on the line 312 causes themodulo-M block 350 to generate the RESIDUE signal, as described above.The RESIDUE value is applied to the LED modulation block 104 whichgenerates the modulation signals M₁(t) and M₂(t), as described above.The RESIDUE value is also applied to the LED demodulation state tableblock 352 which generates a new value for D₁(t) and a new value forD₂(t) for each new RESIDUE value. Thus, 148 values of D₁(t) and D₂(t)are generated for each complete cycle. Because the clock signal isoperating at 46,875 Hz, the modulation signals M₁(t) and M₂(t) and thedemodulation signals D₁(t) and D₂(t) have a fundamental frequency of316.722973 Hz, which, as discussed above, does not correspond to anyharmonic of conventional 50 Hz or 60 Hz power line frequencies.

[0114] The HWΔ (hardware delay factor) value, the HWD (hardwaredistortion factor) value and the SCL (scaling factor) value are foundempirically as follows. First, the ideal values of the hardware delayfactor, the hardware distortion factor and the scale factor are appliedto the Equations 32 and 33 in the LED demodulation state table block 352(i.e., HWΔ=0, HWD=1.414213562, and SCL=2.221441469). To determine theoptimum value of the hardware delay factor, the second modulation signalM₂(t) is set to a constant value of zero (i.e., the infrared LED ismaintained in its OFF state). The red LED pulses are applied as setforth above, and the digital detector output signal from theanalog-to-digital converter is monitored and compared to the modulationsignal M₁(t). The relative delay between the beginning of the modulationsignal M₁(t) and the detection of the beginning of the responsive outputfrom the analog-to-digital converter is the optimum hardware delayfactor (HWΔ) value. In one exemplary embodiment, the optimum value ofthe hardware delay factor is 39.

[0115] After determining the value of the hardware delay factor andapplying it to Equations 32 and 33, the ideal value of the hardwaredistortion factor and the ideal value of the scale factor are applied tothe two equations. Again, with the red LED pulses applied to the red LED106 and no pulses applied to the infrared LED, the value of the hardwaredistortion factor is slowly varied from its ideal value while the DCcomponent of the demodulated infrared signal output on the line 342 ismonitored. The value of the hardware distortion factor is varied untilthe measured DC component is minimized, and the value of the hardwaredistortion factor corresponding to the minimal DC component is selectedas the optimum value for the hardware distortion factor.

[0116] Next, with the value of the hardware delay factor and the valueof the hardware distortion factor set to their respective optimumvalues, as determined above, the value of the scale factor (SCL) isinitially set to 1. Again, with the modulation system generating pulsesonly to the red LED 106, the DC component of the demodulated red signaloutput on the line 340 is measured. In addition, the difference inamplitude between the on state and the off state of the digital detectorsignal from the filter 198 is measured. The ratio of the measuredamplitude difference to the measured DC component of the demodulated redsignal output is selected as the optimum value for the scale factor.

[0117] An exemplary demodulation waveform D₁(t) is illustrated by awaveform 400 in FIG. 11 and an exemplary demodulation waveform D₂(t) isillustrated by a waveform 402 in FIG. 11. The demodulation waveforms inFIG. 11 are illustrated with the hardware delay factor set to 0 in orderto align the waveforms with the modulation waveforms in FIG. 10. Itshould be understood that when the hardware delay factor is non-zero,the demodulation waveforms in FIG. 11 will be shifted in phase withrespect to the modulation waveforms in FIG. 10.

[0118] Although described above in connection with the variation of theamplitude of the first harmonic component of the demodulation signals inorder to minimize the crosstalk, it should be understood that therelative amplitude of the second harmonic component of the demodulationsignals with respect to the amplitude of the fundamental component ofthe demodulation signals is determined by the relationship of theamplitude of the first harmonic component of the modulation signals tothe amplitude of the fundamental component of the modulation signals.The relationship of the amplitude of the first harmonic component of themodulation signals depends in part upon the duty cycles of themodulation signals. If the modulation duty cycles are varied, theamplitude of the first harmonic component of the modulation signalschanges. Thus, the crosstalk may also be minimized by holding theamplitudes of the components of the demodulation signals constant whilevarying the duty cycles of the modulation signals. One skilled in theart will appreciate that other variations in the modulation anddemodulation signals may also be used to minimize the crosstalk betweenthe two output signals.

[0119] A plurality of signals S₁, S₂, S₃ . . . S_(n) can be demodulatedand the crosstalk between signals reduced to a minimum by application ofthe foregoing invention to more than two signals.

[0120] Additional information can advantageously be derived from thedigitized detection signal on the bus 314 and can be used to provideindications regarding the reliability of the demodulated signalsgenerated as described above. In particular, although the present systemis capable of demodulating the Ŝ₁(t) signal and the Ŝ₂(t) signal in thepresence of significant ambient noise from light and other sources, itis possible that the level of the ambient noise is sufficiently high toaffect the demodulated signals. FIGS. 12 and 13 illustrate a time domainmethod and system for determining the ambient noise level, and FIGS. 14and 15 illustrate a frequency domain method and system for determiningthe ambient noise level.

[0121] As illustrated in FIG. 12, the digital detection signal 152 issampled by a sample signal represented by a waveform 500, whichcomprises a plurality of sampling pulses 502. The sampling pulses 502are timed to occur during the intervals between the red pulses 134, 136and the infrared pulses 142, 144 when no red light and no infrared lightshould be detected by the detector 150 (FIG. 1). Thus, any energydetected during the sample intervals is primarily caused by ambientlight and other noise sources. As illustrated, the sampling pulses 502preferably occur at the approximate midpoint of each interval betweenthe red and infrared pulses.

[0122] As illustrated in FIG. 13, the digital detection signal bus 314is provided as an input to a time domain sampler 520. The time domainsampler 520 also receives the RESIDUE signal on the bus 356 as a secondinput. The time domain sampler is responsive to the RESIDUE signal tosample the digital detection signal at times when the value of theRESIDUE signal corresponds to the quiescent times of the red pulses 134,136 and the infrared pulses 142, 144. As described above, the red pulses134, 136 are generated when the RESIDUE signal has values between 0 and36, and the infrared pulses are generated when the RESIDUE signal hasvalues between 74 and 110. Thus, assuming no hardware delay, thesampling pulses 502 are preferably generated, for example, when theRESIDUE signal has a value of 55 and when the RESIDUE signal has a valueof 129, which positions the sampling pulses at the approximate midpointsof the quiescent intervals between the pulses. As discussed above, theactual system has a hardware delay caused by processing times. Thus, ifthe system has a hardware delay factor of, for example, 39, the samplingpulses 502 are shifted in time to occur when the RESIDUE signal has avalue of 94 and a value of 20 (168 modulo₁₄₈). The sample times used bythe time domain sampler 520 are advantageously determined byconfiguration signals received via the digital configuration bus 310,described above. For example, the time domain sampler 520 can beinitially set to sample at RESIDUE signal values of 55 and 129, and thevalue of the hardware delay value factor (HWΔ) communicated by thedigital configuration bus 310 is added to both values to shift thesample to the correct sample interval.

[0123] As illustrated in FIG. 14, a detection signal spectra 550includes the two frequency components corresponding to the fundamentaland first harmonic of the modulation signal at 316.7 Hz and 633.4 Hz,respectively. The spectra 550 further includes the fundamental andmultiple harmonics of the 60 Hz power line frequency. In addition, thespectra 550 includes noise at a multitude of frequencies which may becaused by various sources. One particularly troublesome source of noiseencountered in pulse oximetry systems is an electrocauterization device,which uses a high frequency electrical current to make surgicalincisions and to cauterize the surrounding blood vessels at the sametime. Although primarily high frequency noise sources, such devices alsogenerate significant noise at lower frequencies because of arcing. Whenan electrocauterization device is operated close to a pulse oximeterdetector, the noise generated by the device can overwhelm the signalsgenerated by the pulse oximetry detector. In other words, the noisefloor can be greater than the detectable signal from the pulse oximetrydetector.

[0124] It is desirable to detect when the noise floor is too high sothat the pulse oximetry system can indicate that the demodulated signalsmay not be reliable. In order to determine the level of the noise floor,the present invention samples the spectra 550 to determine the contentof the frequency components detected at frequencies other than thefundamental and harmonic frequencies of the modulation signals. Inparticular, as illustrated by a sample control signal 560 in FIG. 14,the portions of the spectra 550 which do not include the fundamental andharmonics of the modulation signal are sampled. Thus, in the preferredembodiment, the magnitudes of the spectra at 316.7 Hz, 633.4 Hz, 950.1Hz, etc., are not sampled. Furthermore, because a band of frequenciesaround the fundamental and harmonics of the modulation signal alsoinclude significant information caused by the modulation of the redpulses and the infrared pulses by the changes in blood flow during eachcardiac cycle. Thus, as illustrated in FIG. 14, in the preferredembodiment, a band of frequencies surrounding the fundamental andharmonic frequencies of the modulation signals (i.e., the sidebandsdiscussed above) are not included in the samples. For example, a band ofat least ±10 Hz around each of the fundamental and harmonic frequenciesis not included in the samples.

[0125] The intensities at the sampled frequencies are averaged, and anoutput signal is generated which represents the average intensity of thenoise signals. Other portions (not shown) of the digital processingsystem advantageously monitor the average intensity of the noisesignals, and, if the average intensity exceeds a selected thresholdbased upon the size of the measured plethysmograph, then the demodulatedoutput signals from the system are considered as being unreliable andshould not be used.

[0126]FIG. 15 illustrates a preferred embodiment of a system thatdetermines the noise floor, as described above. The system of FIG. 15includes a Fast Fourier Transform block 600 which receives a pluralityof samples from the digitized detector bus 314 and generates atransformed output on a bus 610. The transformed output on the bus 610represents the spectra of the samples. In the preferred embodiment, asufficient number of samples are taken to represent approximately 44milliseconds of data so that at least two cycles of the 60 Hz power areincluded within the samples. For example, approximately 1,024 samplescan be taken during the 44-millisecond interval at a sample rate ofapproximately 23.4 kHz (e.g., one-half the system timing rate). Thespectra for a 44-millisecond interval are provided as inputs to aspectral sampler 620 which eliminates the samples in the ±10 Hz bandsaround the fundamental and harmonic frequencies of the modulationsignals. The output of the spectral sampler 620 is provided on a bus 630and is thereby provided as an input to an averager 640. The averager 640averages the sampled noise spectra which it receives and provides anaveraged output on a bus 650. The averaged output on the bus 650represents the noise floor and is provided to other portions of thedigital processing system where it is compared to the selected thresholdto determine whether the noise floor is excessive. The threshold is notnecessarily fixed, but is dependent on the strength of theplethysmograph, which in turn depends upon the perfusion of blood in thebody portion being measured.

[0127] The embodiment of FIG. 15 can also advantageously be used todetermine whether the ambient noise is primarily at 60 Hz, correspondingto power line frequencies in the United States and Canada, or at 50 Hz,corresponding to power line frequencies in Europe. The foregoingmodulation frequency of 316.7 Hz is selected to avoid the harmonics ofthe 60 Hz power line frequency as well as the 50 Hz power linefrequency. If a significant shift in the power line frequency isdetected such that aliasing of the ambient noise occurs at thefrequencies of interest, then the modulation frequency can be changed todisplace the modulation harmonics farther from the harmonics of thepower line frequency, such as, for example, by changing the 46,875 Hzsampling frequency, or by changing the modulus.

[0128] Pre-Demodulation Decimation

[0129] For convenience, the previous embodiments do not show the signalMF(k) being decimated before demodulation. However, as discussed in moredetail below, the signal MF(k) can advantageously be decimated prior todemodulation. The pre-demodulation decimation technique can reduce thecomputational burden required to perform the demodulation operations,primarily because the decimated sample rate is lower than the original(undecimated) sample rate. Computation can also be reduced because, aswill be seen, the numerical sequences used in the demodulator are, insome circumstances, shorter than the sequences given in Equations 32 and33. Pre-demodulation decimation is a generalization of the previousembodiments and reduces to the previous embodiments when thepre-demodulation decimation rate is one.

[0130]FIG. 16 is a pictorial representation of a system thatincorporates pre-demodulation filtering and decimation. FIG. 16 issimilar to FIG. 3, and like numbers refer to like elements in the twofigures. FIG. 16 shows the first modulator 191 having a signal inputS₁(t) and a modulation input M₁(t). The second modulator 193 has asignal input S₂(t) and a modulation input M₂(t). The pair of signalsS₁(t) and S₂(t) represent the effect of the time-varying volume andscattering components of the blood in a finger (or other body part) onthe red light and the infrared light, respectively, passing through thefinger. The red light signal portion S₁(t) is caused by the variableattenuation of the red light passing through the finger 102 (shown inFIG. 1). The infrared light signal portion S₂(t) is caused by thevariable attenuation of the infrared light passing through the finger102. The outputs of the first and second modulators 191, 193 areprovided to the receiving photodetector 150. The photodetector 150 ismodeled as an adder 194 and an adder 196. The outputs of the first andsecond modulators 191, 193 are provided to the adder 194 to generate acomposite signal M(t) where:

M(t)=S ₁(t)M ₁(t)+S₂(t)M ₂(t).  (34)

[0131] The output signal M(t) from the adder 194 is provided to an adder196 where a signal n(t) is added to the signal M(t). The signal n(t)represents a composite noise signal caused by ambient light (includingDC and harmonics of the power line frequency), electromagnetic pickup,and the like, which are also detected by the photodetector 150. Inaddition, the signal n(t) may also include noise at higher frequenciescaused, for example, by other devices such as electrocauterizationequipment, or the like. The output of the adder 196 is a signalM′(t)=M(t)+n(t) which includes noise components as well as the signalcomponents.

[0132] The M′(t) signal output of the adder 196 (i.e., the output of thedetector 150) is applied to the input of a signal processing block 1600.Within the signal processing block 1600, the signal M′(t) is firstpassed through the amplifier 197 and then through the analog bandpassfilter 198. The analog bandpass filter 198 provides anti-aliasing andremoval of low frequency noise and DC. The filter 198 has a passbandselected to pass signals in the preferred range of 20 Hz to 10,000 Hz.The analog bandpass filter 198 removes a significant portion of thenoise below 20 Hz. The signal components responsive to the blood oxygensaturation are frequency shifted by the operation of the two modulationsignals M₁(t) and M₂(t) and are passed by the analog bandpass filter198.

[0133] In one embodiment, the output of the analog bandpass filter 198is sampled by the analog-to-digital converter 199 and converted thereinto digital signals. In one embodiment, the signals are sampled at 46,875samples per second. The digital signals from the analog-to-digitalconverter 199 are provided as inputs to a lowpass digital filter 1620.Output signals from the digital filter 1620 are provided to a samplerate compression block 1622 that reduces (compresses) the sample rate bya decimation rate R₁. The lowpass digital filter 1620 and sample ratecompressor 1622 together comprise a decimator 1621 (decimation compriseslowpass filtering followed by sample rate compression). The digitalfilter 1620 provides anti-aliasing filtering and the sample ratecompression block 1622 preferably operates at a sampling rate of atleast twice the highest frequency of interest as determined by thedigital filter 1620. In one embodiment, the sample rate compressionblock 1622 reduces the sample rate by a factor of R₁=37, correspondingto the number of samples during the period τ as illustrated in FIG. 10.The output of the sample rate compression block 1622 provides one sampleper time period τ and thus four samples per time period T. The output ofthe sample rate compression block 1622 is a signal MF(k) (where k is adiscrete index) which comprises approximately 1,266 samples per second.

[0134] The signal MF(k) is provided as a first input to a first mixer1624. The signal MF(k) is also provided as a first input to a secondmixer 1626. A first demodulating signal D₁(k) is provided as a secondinput to the first mixer 1624, and a second demodulating signal D₂(k) isprovided as a second input to the second mixer 1626. The output of thefirst mixer 1624 is provided as an input to a first lowpass filter 1630,and the output of the second mixer is provided as an input to a secondlowpass filter 1640. The bandwidths of the lowpass filters 1630, 1640are preferably approximately 10 Hz. The signal MF(k) is also provided asa first input to a noise channel mixer 1628. A noise demodulating signalD₀(k) is provided as a second input to the noise channel mixer 1628. Theoutput of the noise channel mixer 1628 is provided to an input of a lowpass filter 1650. The output of the low pass filter 1650 is provided toa sample rate compression block 1652. The output of the sample ratecompression block 1652 is an estimate of the noise n(t). The output ofthe lowpass filters 1630 is provided to an input of a sample ratecompressor 1632 and the output of the lowpass filter 1640 is provided toan input of a sample rate compressor 1642. The lowpass filter 1630 andthe sample rate compressor 1632 together comprise a decimator 1631. Thelowpass filter 1640 and the sample rate compressor 1642 togethercomprise a decimator 1641.

[0135] The output of the decimator 1631 is a signal Ŝ₁(k), which, asdiscussed below, is an estimate of the signal S₁(k). The output of thedecimator 1641 is a signal Ŝ₂(k), which, as discussed below, is anestimate of the signal S₂(k). As will be shown below, the selection ofthe first demodulating signal D₁(k) and the second demodulating signalD₂(k) in accordance with the present invention can reduce or eliminatethe effects of noise in the two output signals Ŝ₁(k) and Ŝ₂(k) and alsoreduce or eliminate crosstalk between the two signals.

[0136] The decimators 1632, 1642 decimate by a decimation rate R₂. In apreferred embodiment, the decimators 1632, 1642 decimate by a decimationrate R2=20 to a sample rate of, for example, 63.3 Hz to provide adecimated output which can be further processed in accordance with themethods and apparatuses described in the above-referenced patents. Thedecimations which occur in the decimators 1632, 1642 reduce the rate atwhich the output signals Ŝ₁(k) and Ŝ₂(k) need to be processed whilemaintaining the sample rate well above the 10 Hz frequency content ofthe signals of interest. The outputs of the decimators 1632, 1642 areprovided on respective output lines 1634 and 1644.

[0137] Decimating the signal MF(k) prior to demodulation, although notan approximation technique, can be simplified by assuming that eachdesired signal S₁(t) does not change appreciably during each period τ.In many applications it is reasonable to assume that the desired signalsS₁(t) and S₂(t) will not change significantly during the time interval τshown in FIG. 2. One skilled in the art will recognize that a sufficientcondition for this assumption is that the highest significant frequencycomponents in S₁(t) and S₂(t) are much lower than the modulationfrequency. In the pulse-oximetry application the highest frequency ofinterest is typically around 10 Hz, which is far below the 316.7 Hzfundamental of the modulation. Since n(t) is not a desired signal, nosuch assumption is necessary for n(t). Thus, while n(t) may varyerratically over a modulation cycle, the signals S₁(t) and S₂(t) do not.Therefore, it is possible to perform pre-demodulation decimation thathas little effect on S₁(t) and S₂(t) but may shape n(t) into n′(t). Themeasured signal is decimated by a factor R₁=Q (where Q is the number ofsamples in a time period τ) and then demodulated.

[0138] Assuming R₁=Q, then the spectral domain representation of thesignal MF(k) at the output of the sample rate compression block 1622 isgiven by (approximately): $\begin{matrix}\begin{matrix}{{{MF}(f)} = {{\frac{1}{T}{\sum\limits_{n = {- \infty}}^{\infty}\lbrack {{S_{1}( {f - \frac{n}{T}} )} + {( {- 1} )^{n}\quad {S_{2}( {f - \frac{n}{T}} )}}} \rbrack}} +}} \\{{\frac{4}{T}{\sum\limits_{m = {- \infty}}^{\infty}{n^{\prime}( {f - \frac{4\quad m}{T}} )}}}}\end{matrix} & (35)\end{matrix}$

[0139] Since the sample rate compression block 1622 decimates at thesame rate as the number of samples per period τ, the decimation removesany τ dependence in the expression for MF(f). The frequency componentsindexed by m increase four times faster than the frequency componentsindexed by n. This occurs because the modulated signals S₁(t) and S₂(t),which are indexed by n, occur in only one fourth of the samples, but thenoise n(t), which is indexed by m, occurs in every sample.

[0140] The demodulation operation can be performed either in thefrequency or the time domain. A method for frequency domain demodulationof the signal MF(k) can be obtained by rewriting Equation 35 as:

MF(f)= . . . MF ⁻²(f)+MF ⁻¹(f)+MF ₀(f)+MF ₁(f)+MF ₂(f)+ . . .  (36)

[0141] where

MF ⁻²(f)=[S ₁(f)+S ₂(f)]/T

MF ⁻¹(f)=[S ₁(f)−S ₂(f)]/T

MF ₀(f)=[S ₁(f)+S ₂(f)+4n′(f)]/T

MF ₁(f)=[S ₁(f)−S ₂(f)]/T

MF ₂(f)=[S ₁(f)+S ₂(f)]/T

MF ₃(f)=[S ₁(f)−S ₂(f)]/T

MF ₄(f)=[S ₁(f)+S₂(f)+4n′(f)]/T  (37)

[0142] Where n′(k) is the decimated noise signal n(t). Estimates for thesignal S₁(f) can be obtained by shifting the spectra of MF₁(f) andMF₂(f) by −1/T and −2/T, respectively, and then dividing the sum of theresultant by 2. Likewise, S₂(f) can be obtained by dividing thedifference of the resultant spectra by 2. In other words:

Ŝ ₁(f)=MF ₁(f−1/T)+MF ₂(f−2/T)

Ŝ ₂(f)=MF ₁(f−1/T)MF ₂(f−2/T)  (38)

[0143] Demodulation in the time domain is a more elegant method forobtaining S₁(k) and S₂(k). Time domain demodulation is obtained by usingthe frequency shift property of the Fourier transform given by:

F(ω+ω₀)

e ^(−jω) ^(₀) ^(t) f(t)  (39)

[0144] According to Equation 39, the frequency domain terms MF_(i)(f)are related by a time shift in the time domain and this property can beused to generate the demodulation sequences D₀-D₂. A more completedevelopment of this process (for the general case of N channels) isprovided in Equations 42-50 below and in the text accompanying thoseequations. For the present case, where N=2, using equations 42-50 gives:

D ₀(k)=0, 1, 0, 1, . . .

D ₁(k)=1, −0.5, 0, −0.5, . . .

D ₂(k)=0, −0.5, 1, −0.5, . . .  (40)

[0145] The sequences shown in Equation 40 are repeating sequences of thefour values shown. Thus, the demodulation waveforms are no more thanshort repeating sequences of simple coefficients. Since the samplesMF(k) are time domain sequences, demodulation simply involvesmultiplying the samples MF(k) by the sequences in Equation 40. Forexample, the sequence of coefficients D₀(k)=(0, 1, 0, 1, . . . ) isprovided to the multiplier 1628 to demodulate the signal MF(k) andproduce the estimate of n(k). Similarly, the sequence of coefficientsD₁(k)=(1, −0.5, 0, −0.5, . . . ) is provided to the multiplier 1624 todemodulate the signal MF(k) and produce the estimate of S₁(k).

[0146] Multiple Channel Modulation and Demodulation

[0147] The two-channel pre-demodulation decimation technique describedin the previous section can be extended to multi-channel systems havingmore than two desired signals. FIG. 17 illustrates an expansion of thetwo-channel modulator into a multi-channel modulator/demodulator. FIG.17 shows the first modulator 191 and the second modulator 193 as shownin FIG. 16. Further, FIG. 17 shows a third modulator 1701 and an N^(th)modulator 1702. The signal input S₁(t) and a modulation input M₁(t) areprovided to the first modulator 191. The signal input S₂(t) and amodulation input M₂(t) are provided to the second modulator 193. Asignal input S₃(t) and a modulation input M₃(t) are provided to thethird modulator 1701. A signal input S_(N)(t) and a modulation inputM_(N)(t) are provided to the N^(th) modulator 1702.

[0148] The photodetector 150 is modeled as an adder 194 and an adder196. The outputs of the modulators 191, 193, 1701, and 1703 are addedtogether in the adder 194, to generate a composite signal M(t) where:

M(t)=S ₁(t)M ₁(t)+S₂(t)M ₂(t)+S ₃(t)M ₃(t)+ . . . +S _(N)(t)M_(N)(t)  (41)

[0149] The signal M(t) from the adder 194 is provided to the adder 196where the signal M(t) is added to the signal n(t) which represents acomposite noise signal caused by ambient light, electromagnetic pickup,and the like, which are also detected by the photodetector 150. Theoutput of the adder 196 is the signal M′(t)=M(t)+n(t), which includesthe noise components as well as the signal components.

[0150] The M′(t) signal output of the adder 196 (i.e., the output of thedetector 150) is applied to the input of the signal-processing block1700. Within the signal-processing block 1700, the signal M′(t) is firstpassed through an amplifier 197 and then through the analog bandpassfilter 198. The analog bandpass filter 198 provides anti-aliasing andremoval of low frequency noise and DC. The desired signal components inthe signals S_(i)(t) are frequency shifted by the operation of themodulation signals M_(i)(t) and are passed by the analog bandpass filter198.

[0151] The output of the analog bandpass filter 198 is sampled by theanalog-to-digital converter 199 and converted therein to digital signalsand provided to an input of the lowpass digital filter 1620. Outputsignals from the digital filter 1620 are provided to a sample ratecompression block 1622, which reduces the sample rate by a decimationfactor R₁. Together, the digital filter 1620 and the sample ratecompression block 1622 comprise a decimator 1621. The output of thesample rate compression block 1622 is a signal MF(k). The signal MF(k)is provided as: the first input to the first mixer 1624; the first inputto the second mixer 1626; a first input to a third mixer 1710; a firstinput to an N'th mixer 1712; and a first input to a noise channel mixer1713. A first demodulating signal D₁(k) is provided as a second input tothe first mixer 1624. A second demodulating signal D₂(k) is provided asa second input to the second mixer 1626. A third demodulating signalD₃(k) is provided to the third mixer 1710. A fourth demodulating signalD₀(k) is provided to the N^(th) mixer 1712. A noise demodulating signalD₀(k) is provided to the noise channel mixer 1713. The outputs of themixers 1624, 1626, 1710, 1712, and 1713 are provided as respectiveinputs of the lowpass filters 1630, 1640, 1720, 1730, and 1740. Theoutputs of the lowpass filters 1630, 1640, 1720, 1730, and 1740 areprovided as respective inputs of the decimators 1632, 1642, 1721, 1731and 1741. Each of the decimators 1632, 1642, 1721, 1731 and 1741 reducesthe sample rate by a decimation rate R₂.

[0152] The output of the sample rate compressor 1632 is a signal Ŝ₁(k),which, as discussed below, is an estimate of the signal S₁(k). Likewise,the output of the sample rate compressor 1642 is an estimate of S₂(t),the output of the sample rate compressor 1721 is an estimate of thesignal S₃(t), the output of the sample rate compressor 1731 is anestimate of the signal S_(N)(t), and the output of the sample ratecompressor 1741 is an estimate of the signal n(t).

[0153] As will be shown below, the selection of the demodulating signalsD_(i)(t) for i=0 . . . N in accordance with the present invention cansubstantially reduce or eliminate the effects of noise in the outputsignals Ŝ_(i)(k) and n(k), and can also substantially reduce oreliminate crosstalk between the signals.

[0154] As shown in FIG. 17, a set of N+1 signals S_(i)[k] i=1 . . . N,and n(k) are sampled at a rate T/QN, where T is a modulation period. Forsimplicity, the decimation rate R₁ is assumed to be the same as thefactor Q. The assumption that R₁=Q is not a necessary assumption, butrather is used here to simplify the mathematics. The signals arecombined according to the formula:

S(k)=M ₁(k)S ₁(k)+M ₂(k)S ₂(k)+M ₃(k)S ₃(k)+ . . . +M _(N)(k)S_(N)(k)+n(k)  (42)

[0155] Using the symbol * to denote the convolution operator, the termsM_(i)(k) are given by:

M ₁(k)=Δ(2Nt/T)*P ₁(t)|_(t=kT/QN)

M ₂(k)=Δ(2Nt/T)*P ₂(t)|_(t=kT/QN)

M _(N)(k)=Δ(2Nt/T)*P _(N)(t)|_(t=kT/QN)

[0156] where $\begin{matrix}{{\Delta (x)} = \{ \begin{matrix}1 & {{{if}\quad {x}} \leq 0.5} \\0 & {otherwise}\end{matrix} } & (44) \\{and} & \quad \\{{P_{1}(t)} = {\sum\limits_{n = {- \infty}}^{\infty}{\delta ( {t - {nT}} )}}} & (45)\end{matrix}$

[0157] (where δ(k) is the Kröneker delta function, which is 1 for k=0,and 0 for all other values of k), and $\begin{matrix}\begin{matrix}{{P_{i}(t)} = {P_{1}( {t - \frac{( {i - 1} )T}{N}} )}} & {{{for}\quad i} = {2\quad \ldots \quad N}}\end{matrix} & (46)\end{matrix}$

[0158] After the pre-demodulation and sample rate compression stage1622, which decimates by a factor Q, the signal in the frequency domainis given approximately by $\begin{matrix}\begin{matrix}{{{MF}(f)} = {\frac{1}{T}{\sum\limits_{n = {- \infty}}^{\infty}\lbrack {{S_{1}( {f - \frac{n}{T}} )} + {\xi_{1}^{n}{S_{2}( {f - \frac{2n}{T}} )}} + \cdots +} }}} \\{ {\xi_{N - 1}^{n}{S_{N}( {f - \frac{Nn}{T}} )}} \rbrack + {\frac{2N}{T}{\sum\limits_{m = {- \infty}}^{\infty}{S_{n}( {f - \frac{2{Nm}}{T}} )}}}}\end{matrix} & (47) \\{where} & \quad \\{\xi_{k}^{n} = ^{\frac{j\quad 2\quad \pi \quad {kn}}{N}}} & (48)\end{matrix}$

[0159] The demodulator sequences are then given by: $\begin{matrix}{{{D_{0}(k)} = \frac{( {1 - ( {- 1} )^{k}} )}{2}}{{D_{1}(k)} = {{P_{1}(k)} - \frac{D_{o}(k)}{N}}}{{D_{2}(k)} = {{P_{2}(k)} - \frac{D_{o}(k)}{N}}}\quad \vdots {{D_{N}(k)} = {{P_{N}(k)} - \frac{D_{o}(k)}{N}}}{where}} & (49) \\{{P_{1}(k)} = {{P_{1}(t)}{_{t = \frac{kT}{2N}}{,{{P_{2}(k)} = {{P_{2}(t)}{_{t = \frac{kT}{2N}}{,\cdots \quad,{{P_{N}(k)} = {{P_{N}(t)}_{t = \frac{kT}{2N}}}}}}}}}}}} & (50)\end{matrix}$

[0160] The post demodulation lowpass filters 1630, 1640, 1720, 1730 and1740, and the post demodulation sample rate compression stages 1632,1642, 1721, 1731 and 1741 suppress high frequency artifacts which areproduced by the modulation/demodulation process. Note that Equation 49reduces to Equation 40 for N=2.

[0161] Adaptive Demodulation

[0162] The multi-channel pre-demodulation decimation technique describedin the previous section can be extended to an adaptive multi-channelsystem having an adjustable pre-demodulation decimation rate and anadjustable post-demodulation decimation rate. FIG. 18 illustrates anexpansion of the multi-channel modulator into a adaptive multi-channelmodulator/demodulator 1800. FIG. 18 shows the first modulator 191 andthe N^(th) modulator 1702 as shown in FIG. 17. The signal input S₁(t)and a modulation input M₁(t) are provided to the first modulator 191. Asignal input S_(N)(t) and a modulation input M_(N)(t) are provided tothe N^(th) modulator 1702.

[0163] The photodetector 150 is modeled as an adder 194 and an adder196. The outputs of the modulators 191, 193, 1701, and 1703 are addedtogether in the adder 194, to generate a composite signal M(t) where:

M(t)=S ₁(t)M ₁(t)+ . . . +S _(N)(t)M _(N)(t)  (51)

[0164] The signal M(t) from the adder 194 is provided to the adder 196where the signal M(t) is added to the signal n(t) which represents acomposite noise signal caused by ambient light, electromagnetic pickup,and the like, which are also detected by the photodetector 150. Theoutput of the adder 196 is the signal M′(t) M(t)+n(t), which includesnoise components as well as the signal components.

[0165] The M′(t) signal output of the adder 196 (i.e., the output of thedetector 150) is applied to the input of the signal-processing block1800. Within the signal-processing block 1800, the signal M′(t) is firstpassed through the amplifier 197 and then through the analog bandpassfilter 198. The analog bandpass filter 198 provides anti-aliasing andremoval of low frequency noise and DC. The desired signal components inthe signals S_(i)(t) are frequency shifted by the operation of themodulation signals M_(i)(t) and are passed by the analog bandpass filter198.

[0166] The output of the analog bandpass filter 198 is sampled by theanalog-to-digital converter 199 and converted therein to digital signalsand provided to an input of a decimation block 1820. The adaptivedecimation block 1820 comprises a digital lowpass filter and a samplerate compressor that reduces the sample rate by the decimation rate R₁.The filter coefficients and decimation rate R₁ are provided to a controlinput of the adaptive decimation block 1820 by an output of an adaptivealgorithm block 1850. Equation 35 assumes that the decimation rate R₁ isequal to Q. However, in general, the value of Q may be different thanthe decimation rate R₁. The output of the adaptive decimation block 1820is a signal MF(k).

[0167] The signal MF(k) is provided to the first input of the firstmixer 1624, to the first input of the N'th mixer 1712, and to the firstinput of the noise channel mixer 1713. A first demodulating signal D₁(k)is provided to a second input of the first mixer 1624 from a signalgenerator 1841. The fourth demodulating signal D_(N)(k) is provided tothe N^(th) mixer 1712 from an output of a signal generator 1831. Thenoise demodulating signal D₀(k) is provided to the noise channel mixer1713 from an output of a signal generator 1832. A control input to eachof the signal generators 1831, 1832, and 1841 is provided by the outputof the adaptive algorithm 1850. In yet another embodiment, the adaptivealgorithm 1850 may also be controlled by other signal processingelements downstream of the signal processor 1800.

[0168] The outputs of the mixers 1713, 1624, and 1712 are provided asrespective inputs to adaptive decimation blocks 1840, 1830, and 1834respectively. Each of the adaptive decimation blocks 1840, 1830, and1834 has a control input provided by the output of the adaptivealgorithm block 1850. The output of the adaptive decimation block 1840is an estimate of the signal n(t) and it is provided to an input of theadaptive algorithm block 1850. In an alternate embodiment, the signalestimates Ŝ_(i)(k) are also provided to the adaptive algorithm block1850.

[0169] An output of the decimator 1830 is a signal Ŝ₁(k), which, asdiscussed above, is an estimate of the signal S₁(k). Likewise, theoutput of the decimation block 1834 is an estimate of the signalS_(N)(t). As shown above, the selection of the demodulating signalsD_(i)(t) for i=0 . . . N in accordance with the present inventionsubstantially reduces or eliminates the effects of noise in the outputsignals Ŝ_(i)(k) and n(k), and also substantially reduces or eliminatescrosstalk between the signals.

[0170] As shown in FIG. 18, a set of N+1 signals S_(i)[k] i=1 . . . N,and n(k) are sampled at a rate T/QN, where T is a modulation period, andR₁ is the decimation rate of the decimation block 1820. The signals arecombined according to the formula:

S(k)=M ₁(k)S ₁(k)+ . . . +M _(N)(k)S _(N)(k)+n(k)  (52)

[0171] Each of the adaptive decimators 1820, 1840, 1830, and 1834comprises a digital lowpass filter and a sample rate compressor. Thecharacteristics of the digital lowpass filters (e.g., the number offilter coefficients and values of the filter coefficients) and thesample rate compression factor of each adaptive decimator is provided toa control input of the adaptive decimator. The control inputs are drivenby an adaptive algorithm 1850. The signal generators 1831, 1832 and 1841generate the demodulation sequences for the demodulators 1624, 1712, and1713 respectively. The demodulation sequences produced by the signalgenerators 1831, 1832 and 1841 are controlled by the adaptive algorithm1850.

[0172] The adaptive algorithm adjusts the pre-demodulation decimationrate R₁ (in the adaptive demodulator 1820), and the post-demodulationdecimation rate R₂ (in the adaptive demodulators 1830, 1834 and 1840)according to the noise in the noise estimate {circumflex over (n)}(k)1746 and (optionally) according to the signals Ŝ_(i)(k). The productR₁R₂ is the total decimation rate from the signal S(k) at the output ofthe A/D converter 199 to the signals Ŝ_(i)(k) at the output of thesignal processing block 1800. The adaptive algorithm may adjust R₁ andR₂ such that the product R₁R₂ varies, or the adaptive algorithm mayadjust R₁ and R₂ such that the product R₁R₂ is substantially constant.Typically, the adaptive algorithm will keep the R₁R₂ product constant sothat the signal processing blocks downstream of the signal processor1800 will operate at a substantially constant sample rate.

[0173] Typically, each of the signal generators 1841, 1831 and 1832generates a repeating sequence of numbers. The number of elements in thesequence is a function of the decimation factor R₁. As discussed abovein connection with FIG. 3, when R₁=1, there are preferably 148 values ineach demodulation sequence. As discussed above in connection with FIG.17, when R₁=37, there are preferably only 4 values in the demodulationsequences.

[0174] The adaptive algorithm selects R₁, R₂, and the filter transferfunctions in the adaptive decimators 1820, 1830, 1834, and 1840 toimprove the quality of the output signals Ŝ_(i)(k). For example, in highambient noise environments, the higher order harmonics of the outputsignals are often contaminated by ambient noise (as discussed inconnection with FIGS. 14 and 20). Thus, the higher order harmonics arepreferably not demodulated when ambient noise is present. To avoiddemodulation of the higher order harmonics the adaptive demodulator 1850can set R₁=1 and R₂=37, and thereby demodulate according to the methoddescribed in connection with FIGS. 3-14. Alternatively, the adaptivedemodulator 1850 can set R₁=37, set R₂=1, and set the transfer functionof the lowpass filter in the adaptive decimator 1820 to provide a veryfast rolloff (thereby filtering out the higher order harmonics).

[0175] Conversely, in low ambient noise environments, the higher orderharmonics of the output signal are less contaminated by ambient noise,and thus the higher order harmonics may be demodulated. In oneembodiment, to demodulate the higher order harmonics, the adaptivedemodulator 1850 can set R₁=37 and set R₂=1, to demodulate according tothe method described in connection with FIG. 17. This is especiallyadvantageous when perfusion is low, because, when perfusion is low theoutput signals Ŝ_(i)(k) are typically very weak and are contaminated byrandom noise. Demodulating more of the higher order harmonics increasesthe signal-to-noise ratio because it adds the harmonics (which arecorrelated) to the output signals, and tends to average out the noise(which is uncorrelated). Thus, the signal strength increases, and thenoise is reduced.

[0176] One skilled in the art will recognize that the examples in thepreceding two paragraphs are merely two points on a continuum and thatthe adaptive algorithm 1850 can generate many desirable solutions on thecontinuum.

[0177] Ambient Light Rejection

[0178] In the pulse oximeter, one of the major contributors to the noisesignal n(t) is ambient light that is detected by the photodetector 150.One aspect of the present invention advantageously provides a method forchoosing the modulation sampling rate f_(s) and the factor Q so that theeffects of ambient light can be removed by the post demodulationfiltering and decimation stages. Note that Q is the number of samplesduring the on period (i.e., modulation signal sample turn on time Q) andis preferably also the decimation rate R₁ for the pre-demodulationsample rate compressor 1622 (in general the values of Q and R₁ may bedifferent). The particular embodiment described by Equation 35 assumesthat the value Q is also used as decimation rate R₁ for thepre-demodulation decimator 1820.

[0179] In the system shown in FIGS. 3 and 16, which demodulates twoharmonics, the period of a modulation cycle is given by:

T=4Q/f _(s)  (53)

[0180] where f_(s) is the sample rate. Defining the two line equations$\begin{matrix}{{{y( {f_{a},n} )} = {{nf}_{a} - \frac{1}{T}}}{{z( {f_{a},n} )} = {{nf}_{a} - \frac{2}{T}}}} & (54)\end{matrix}$

[0181] where

f=line frequencies of concern

n=line frequency harmonic numbers of concern  (55)

[0182] then the effects due to ambient light will be minimized when

|y(f _(a) , n)|≧SBF

|z(f _(a) , n)|≧SBF  (56)

[0183] where SBF is the stop band frequency of the post demodulation anddecimation stages (e.g., the 10 Hz lowpass filter 1630 and the samplerate compressor 1632, etc.).

[0184]FIG. 19 is a flowchart showing a method for selecting f_(s) and Q.The method begins at a process block 1902 wherein the ambient lightfrequencies f_(s) and important harmonic components n are identified.Important harmonics are defined as those harmonics that will degradesystem performance below acceptable levels when detected by the detector150. The process then advances from the process block 1902 to a processblock 1904. In the process block 1904, the values of f_(a) and nidentified in the process block 1902 are used in conjunction withEquation 54 to identify a collection of acceptable values of T. Uponcompletion of the process block 1904, the process advances to a processblock 1806. In the process 1906, suitable values of f_(s) and Q arechosen using the values of T obtained in the process block 1904 and theequation T=4Q/f_(s). One skilled in the art will recognize that, since Tis proportional to the ratio of Q/f_(s), knowing T will not uniquelydetermine either f_(s) or Q.

[0185] For example, given power line frequencies of 50±1 Hz and 60±1 Hzthen the range of f_(a) is given by approximately the union of theinterval 49-51 Hz and the interval 59-61 Hz, which can be expressedmathematically as:

f _(a)≈[49,51]∪[59,61]  (57)

[0186] Assuming that all harmonics up to the 18^(th) harmonic are to besuppressed, then n=1 . . . 18. In a preferred embodiment, using thesevalues for f_(a) and n, application of the method in FIG. 19 results inf_(s)=46,875 Hz and acceptable Q values of 37 and 41.

[0187] The process leading to Equation 57 is illustrated graphically byFIG. 20, where the harmonics of the ambient light frequency f_(a) (inHz) are plotted versus the plethysmograph signal frequency (also in Hz).FIG. 20 has an x-axis showing the ambient light frequency from 44 Hz to64 Hz. The ambient light frequency will usually correspond to thefrequency of the power lines, which is nominally 60 Hz (in the U.S.) and50 Hz (outside the U.S.). However, power line frequency regulationtypically varies somewhat, and thus FIG. 20 shows frequencies above andbelow the nominal frequencies.

[0188]FIG. 20 also shows a y-axis showing the plethysmograph signalfrequency from −10 Hz to 10 Hz. One skilled in the art will recognizethat negative frequencies occur in the mathematics described above. Inparticular, a signal that is modulated from baseband up to some carrierfrequency will exhibit two sidebands, a sideband above the carrierfrequency corresponding to the frequency of the baseband signal, and asideband below the carrier frequency corresponding to the negative ofthe baseband frequency. Thus, when dealing with modulation anddemodulation, it is convenient to deal with positive and negativefrequencies.

[0189]FIG. 20 also shows harmonic lines corresponding to the 5^(th),6^(th), 7^(th), 10^(th), 11^(th), 12^(th), 13^(th), and 14^(th)harmonics of the ambient light frequency. The harmonic lines correspondto the harmonics produced in the plethysmograph signal by thedemodulation (mixing down) of harmonics of the power line frequency. Thelines in FIG. 20 are calculated using Equation 54 for 1/T=316.72 Hz.Some of the harmonic lines correspond to y(f_(a),n) and some correspondto z(f_(a),n) from Equation 54. Harmonic lines that are not shown (e.g.,the line corresponding to the 8^(th) harmonic) fall outside thedisplayed limits of the x-axis and y-axis.

[0190]FIG. 20 can be used to determine the stop band frequencies asshown in Equation 56. For example, the harmonic lines in FIG. 20 showthat for an ambient light frequency of 49 Hz, the 13^(th) harmonic ofthe ambient light frequency will appear in the plethysmograph signal atapproximately 3 Hz. Thus, FIG. 20 shows that for plethysmographbandwidth of 10 Hz, none of the first 14 harmonics of the ambient lightwill appear in the plethysmograph signal for ambient light frequenciesbetween approximately 61.2 Hz and approximately 58.5 Hz, which isconsistent with Equation 57. The first ambient harmonics that do appearfor a plethysmograph bandwidth of 10 Hz are the 5^(th) harmonic and the11^(th) harmonic.

[0191] Other Embodiments

[0192] In the preferred embodiment of the present invention, thehardware described above is implemented in a digital signal processorand associated circuitry. The LED modulation block 104 and the LEDdemodulation state table block 352 comprise algorithms implemented byprogram code executed by the digital signal processor. In addition, theconfiguration variables, such as for example, the hardware delay value,the hardware distortion value and the hardware scale value are providedas inputs to the digital signal processor when it is set up. Forexample, the main operating program of the digital signal processor maybe stored in non-volatile ROM or PROM, and the variables may be storedin flash memory during a setup procedure. Techniques for communicatingto and from a digital signal processor during such setup procedures arewell known to persons of skill in the art, and will not be described indetail herein. For example, the configuration bus 310, discussed above,represents a communication path to the flash memory during such a setupprocedure. The data provided to the configuration bus 310 may beprovided by a system operator (not shown) or the data may be providedfrom look-up tables (not shown) maintained for different embodiments ofthe LEDs 106, 108 and the detector 150.

[0193] Although described above in connection with a pulse oximetrysystem wherein a parameter to be measured is the attenuation of red andinfrared light passing through a portion of a subject's body, it shouldbe understood that the method and apparatus described herein can also beused for other measurements where two or more signals are passed througha system to be analyzed. In particular, the present invention can beused to demodulate two combined parametric signals responsive to thesystem to be analyzed where the two parametric signals have apredetermined timing relationship between them, as described herein.

[0194] One skilled in the art will recognize that the lowpass filtersprovided in connection with the decimation blocks may provide otherfilter functions in addition to lowpass filtering. Thus, for example,the lowpass filters 1620, 1622, 1630, 1640, 1650, 1720, 1730, and 1740,and the decimators 1820, 1830, 1834, and 1840 may provide other filterfunctions (in addition to lowpass filtering) such as, for example,bandpass filtering, bandstop filtering, etc. Moreover, thepost-demodulation decimation rate R₂ need not be the same for eachoutput channel. Thus, for example, in FIG. 18, the decimator 1840 mayhave a first decimation rate R₂=r₁ while the decimators 1830 and 1834have a second decimation rate R₂=r₂.

[0195] Although described above in connection with a particularembodiment of the present invention, it should be understood thedescription of the embodiment is illustrative of the invention and arenot intended to be limiting. Various modifications and applications mayoccur to those skilled in the art without departing from the true spiritand scope of the invention as defined in the appended claims.

What is claimed is:
 1. An apparatus for measuring blood oxygenation in asubject, said apparatus comprising: a first signal source which appliesa first input signal during a first time interval; a second signalsource which applies a second input signal during a second timeinterval; a detector which detects a first parametric signal responsiveto said first input signal passing through a portion of said subjecthaving blood therein and which detects a second parametric signalresponsive to said second input signal passing through said portion ofsaid subject, said detector generating a detector output signalresponsive to said first and second parametric signals; and a signalprocessor which receives said detector output signal, said signalprocessor demodulating said detector output signal by applying a firstdemodulation signal to a signal responsive to said detector outputsignal to generate a first output signal responsive to said firstparametric signal and applying a second demodulation signal to saidsignal responsive to said detector output signal to generate a secondoutput signal responsive to said second parametric signal, each of saidfirst demodulation signal and said second demodulation signal comprisingat least a first component having a first frequency and a firstamplitude and a second component having a second frequency and a secondamplitude, said second frequency being a harmonic of said firstfrequency, said second amplitude selected to be related to said firstamplitude to minimize crosstalk from said first parametric signal tosaid second output signal and to minimize crosstalk from said secondparametric signal to said first output signal.
 2. The method of claim 1,wherein the second amplitude is determined by turning off one of thefirst and second signal sources and measuring the crosstalk between oneof the parametric signals and the non-corresponding output signal whilevarying the second amplitude and selecting a second amplitude whichminimizes the measured crosstalk.
 3. A method of minimizing crosstalkbetween two signals generated by applying a first pulse and a secondpulse to measure a parameter, wherein said first pulse and said secondpulse are applied periodically at a first repetition rate defining aperiod, and wherein said first pulse is generated during a firstinterval in each period and said second pulse is generated during asecond interval in each period, said second interval spaced apart fromsaid first interval, said first and second pulses producing first andsecond parametric signals responsive to said parameter, said first andsecond parametric signals being received by a single detector whichoutputs a composite signal responsive to said first and secondparametric signals, said method comprising the steps of: applying afirst demodulation signal to said composite signal to generate a firstdemodulated output signal, said first demodulation signal comprising atleast a first component having a first frequency corresponding to saidfirst repetition rate and having a first amplitude, said firstdemodulation signal further comprising a second component having asecond frequency which is a harmonic of said first frequency and havinga second amplitude which has a selected proportional relationship tosaid first amplitude; applying a second demodulation signal to saidcomposite signal to generate a second demodulated output signal, saidsecond demodulation signal comprising said first component at said firstfrequency and said first amplitude and comprising said second componentat said second frequency and said second amplitude, at least one of saidfirst and second components of said second demodulation signal having aselected phase difference with respect to the corresponding one of saidfirst and second components of said first demodulation signal; andlowpass filtering said first demodulated output signal to generate afirst recovered output signal responsive to said first parametricsignal; and lowpass filtering said second demodulated output signal togenerate a second recovered output signal responsive to said secondparametric signal.
 4. The method as defined in claim 3, wherein saidselected phase difference is π.
 5. The method as defined in claim 3,wherein: said first pulse and said second pulse are generallyrectangular pulses having a duty cycle, and wherein said rectangularpulses comprise a plurality of sinusoidal components including afundamental component corresponding to said first frequency and a firstharmonic component corresponding to said second frequency, saidfundamental component having a fundamental component amplitude and saidfirst harmonic component having a first harmonic component amplitude,said first harmonic component amplitude related to said harmoniccomponent amplitude by a first proportionality value; and said secondamplitude of said second component of said first demodulation signal isrelated to said first amplitude of said first component of said firstdemodulation signal by a second proportionality value which isapproximately the inverse of said first proportionality value.
 6. Themethod as defined in claim 3, further including the steps of: samplingsaid composite signal when neither said first pulse nor said secondpulse is active to obtain a sampled signal; and measuring said sampledsignal to determine a noise level of said parametric signals.
 7. Themethod as defined in claim 3, further including the steps of: performinga transform on said composite signal to generate a spectra of saidcomposite signal; sampling said spectra at a plurality of frequenciesother than at predetermined ranges of frequencies around said firstfrequency and around harmonics of said first frequency; determining anaverage of the magnitudes of said sampled plurality of frequencies; andcomparing said average to a selected threshold to determine whether theaverage magnitude exceeds said selected threshold.
 8. A method ofdemodulating a composite signal generated by applying first and secondperiodic pulses of electromagnetic energy to a system having a parameterto be measured and by receiving signals responsive to saidelectromagnetic energy after having passed through said system and beingaffected by said parameter being measured, said signals received as acomposite signal having components responsive to said first and secondpulses, said method comprising the steps of: applying a firstdemodulation signal to said composite signal to generate a firstdemodulated signal, said first demodulation signal comprising a firstcomponent having a first frequency corresponding to a repetitionfrequency of said first and second pulses and comprising a secondcomponent having a frequency which is a harmonic of said firstfrequency, said first component having a first amplitude and said secondcomponent having a second amplitude, said second amplitude having apredetermined relationship to said first amplitude, said predeterminedrelationship selected to cause said first demodulated signal to have lowfrequency components responsive only to said first pulse; and lowpassfiltering said first demodulated signal to generate a first outputsignal, said first output signal varying in response to an effect ofsaid parameter on the electromagnetic energy received from said firstpulse.
 9. The method as defined in claim 8, further including the stepsof: applying a second demodulation signal to said composite signal togenerate a second demodulated signal, said second demodulation signalhaving first and second components corresponding to said first andsecond components of said first demodulation signal, at least one ofsaid first and second components of said second demodulation signalhaving a selected phase relationship with the corresponding one of saidfirst and second components of said first demodulation signal; andlowpass filtering said second demodulated signal to generate a secondoutput signal, said second output signal varying in response to aneffect of said parameter on the electromagnetic energy received fromsaid second pulse.
 10. The method as defined in claim 9, wherein saidselected phase relationship is a π phase difference.
 11. A pulseoximetry system, comprising: a modulation signal generator, saidmodulation signal generator generating a first modulation signalcomprising a first pulse which repeats at a first repetition frequency,said first pulse having a duty cycle of less than 50%, said modulationsignal generator generating a second modulation signal comprising asecond pulse which also repeats at said first repetition frequency, saidsecond pulse having a duty cycle of less than 50%, said second pulseoccurring at non-overlapping times with respect to said first pulse,said first and second pulses comprising a plurality of componentswherein a first component has a frequency corresponding to saidrepetition frequency and a second component has a second frequencycorresponding to twice said first frequency, said second componenthaving an amplitude which has a first predetermined relationship to anamplitude of said first component; a first transmitter which emitselectromagnetic energy at a first wavelength in response to said firstpulse; a second transmitter which emits electromagnetic energy at asecond wavelength in response to said second pulse; a detector whichreceives electromagnetic energy at said first and second wavelengthsafter passing through a portion of a subject and which generates adetector output signal responsive to the received electromagneticenergy, said detector output signal including a signal componentresponsive to attenuation of said electromagnetic energy at said firstwavelength and a signal component responsive to attenuation of saidelectromagnetic energy at said second wavelength; a first demodulatorwhich multiplies said detector signal by a first demodulation signal andgenerates a first demodulated output signal, said first demodulationsignal comprising a first component having said first frequency andhaving a first amplitude and comprising a second component having saidsecond frequency and having a second amplitude, said second amplitudehaving a second predetermined relationship to said first amplitude whichsecond predetermined relationship is inversely proportional to saidfirst predetermined relationship; and a second demodulator whichmultiplies said detector signal by a second demodulation signal andgenerates a second demodulated output signal, said second demodulationsignal comprising a first component having said first frequency andhaving said first amplitude and comprising a second component havingsaid second frequency and having said second amplitude, at least onecomponent of said second demodulation signal having a selected phaserelationship with a corresponding one component of said firstdemodulation signal.
 12. The method as defined in claim 11, wherein saidselected phase relationship is a π phase difference.
 13. A method ofminimizing crosstalk between two signals generated by applying a firstpulse and a second pulse to measure a parameter, wherein said firstpulse and said second pulse are applied periodically at a firstrepetition rate defining a period, and wherein said first pulse isgenerated during a first interval in each period and said second pulseis generated during a second interval in each period, said secondinterval spaced apart from said first interval, said first and secondpulses producing first and second parametric signals responsive to saidparameter, said first and second parametric signals being received by asingle detector which outputs a composite signal responsive to saidfirst and second parametric signals, said method comprising the stepsof: providing said composite signal to an analog to digital converter toproduce a sequence of digital values; decimating said sequence ofdigital values to produce a decimated sequence of digital values;applying a first sequence of demodulation coefficients to said decimatedsequence of digital values to generate a first demodulated outputsignal; applying a second sequence of demodulation coefficients to saiddecimated sequence of digital values to generate a second demodulatedoutput signal; lowpass filtering said first demodulated output signal togenerate a first recovered output signal responsive to said firstparametric signal; and lowpass filtering said second demodulated outputsignal to generate a second recovered output signal responsive to saidsecond parametric signal.
 14. The method as defined in claim 13, furthercomprising the steps of: applying a third sequence of demodulationcoefficients to said decimated sequence of digital values to generate athird demodulated output signal; and lowpass filtering said thirddemodulated output signal to generate a recovered output signalresponsive to noise detected by said detector.
 15. A method ofdemodulating a composite signal generated by applying a plurality ofperiodic pulse trains of electromagnetic energy to a system having aparameter to be measured and by receiving signals responsive to saidelectromagnetic energy after having passed through said system and beingaffected by said parameter being measured, said signals received as acomposite signal having components responsive to said plurality ofperiodic pulse trains, each of said components corresponding to one eachof said pulse trains, said method comprising the steps of: sampling saidcomposite signal using an analog to digital converter to produce asequence of digital values; decimating said sequence of digital valuesto produce a decimated sequence of values; providing said decimatedsequence of values to a first input of each of plurality of multipliers;providing a plurality of sequences of demodulation coefficients to asecond input of each of said multipliers such that each multiplier isprovided a unique sequence of demodulation coefficients; and lowpassfiltering the output of each of said multipliers such that the outputsignal of each of said lowpass filters approximately corresponds to oneeach of said components.
 16. A pulse oximetry system, comprising: amodulation signal generator, said modulation signal generator generatinga first modulation signal comprising a first pulse which repeats at afirst repetition frequency, said first pulse having a duty cycle of lessthan 50%, said modulation signal generator generating a secondmodulation signal comprising a second pulse which also repeats at saidfirst repetition frequency, said second pulse having said duty cycle ofless than 50%, said second pulse occurring at non-overlapping times withrespect to said first pulse; a first transmitter which emitselectromagnetic energy at a first wavelength in response to said firstpulse; a second transmitter which emits electromagnetic energy at asecond wavelength in response to said second pulse; a detector whichreceives electromagnetic energy at said first and second wavelengthsafter passing through a portion of a subject and which generates adetector output signal responsive to the received electromagneticenergy, said detector output signal including a signal componentresponsive to attenuation of said electromagnetic energy at said firstwavelength and a signal component responsive to attenuation of saidelectromagnetic energy at said second wavelength; a sampling analog todigital converter which converts said detector output into a sequence ofdigital values; a decimator which decimates said sequence of digitalvalues to produce a decimated sequence; a first demodulator whichmultiplies said decimated sequence by a demodulation sequence andgenerates a first demodulated output signal; and a second demodulatorwhich multiplies said decimated sequence by a second demodulationsequence and generates a second demodulated output signal.
 17. Theapparatus of claim 16 further comprising a third demodulator whichmultiplies said decimated sequence by a third demodulation sequence andgenerates a third demodulated output signal corresponding to noiseproduced by said detector.
 18. The apparatus of claim 16 wherein saiddecimator has a decimation rate equal to the number of said samplesproduced during said duty cycle.
 19. In a system that includes: asampling frequency generator generating a sampling frequency f_(s); adetector, a modulation signal generator, said modulation signalgenerator generating a sequence of pulses having a pulse repetitionfrequency, said pulses having a duty cycle of Q sample periods of saidsampling frequency f_(s); a transmitter which emits electromagneticenergy at a wavelength in response to said pulse, said detectorreceiving said electromagnetic energy to generate a detector outputsignal responsive to the received electromagnetic energy, said detectoroutput signal including noise caused by ambient electromagnetic energydetected by said detector; a digital to analog converter which producesdigital samples of said detector output signal at said samplingfrequency; and a demodulator which demodulates said digital samples toproduce a desired output signal, a method for minimizing said noise dueto ambient electromagnetic energy in said desired output signalcomprising the steps of: identifying all undesired frequency componentsof said ambient electromagnetic energy detected by said detector; usingsaid undesired frequency components to compute a set of acceptablemodulation cycle times T; and using said acceptable modulation cycletimes to select said f_(s) and said Q using an equation T=4Q/f_(s). 20.A method for demodulating a multi-channel composite signal comprisingthe acts of: generating a demodulation sequence for a selected channel;providing said demodulation sequence to a first input of a demodulator;providing a sampled composite signal to a second input of saiddemodulator.
 21. The method of claim 20, further comprising the act ofdecimating an output of said demodulator by a decimation factor R. 22.The method of claim 21, wherein said act of decimating includes the actsof: lowpass filtering; and sample rate compressing.
 23. The method ofclaim 20, further comprising the act of decimating said sampledcomposite signal before providing said sampled composite signal to saidsecond input of said demodulator.
 24. The method of claim 23, whereinsaid act of decimating includes the acts of lowpass filtering and samplerate compressing.
 25. The method of claim 24, wherein said acts oflowpass filtering and sample rate compressing are controlled by anadaptive algorithm.
 26. The method of claim 20, further comprising theacts of: decimating in a first decimator said sampled composite signalbefore providing said sampled composite signal to said second input ofsaid demodulator, wherein said first decimator has a first lowpassfilter transfer function and a first decimation rate; and decimating ina second decimator an output of said demodulator, wherein said seconddecimator has a second lowpass filter transfer function and a seconddecimation rate.
 27. The method of claim 26, further comprising the actof controlling said first lowpass filter, said second lowpass filter,said first decimation rate, and said second decimation rate by anadaptive algorithm.
 28. The method of claim 26, wherein a product ofsaid first decimation rate and said second decimation rate issubstantially constant.